Systems and methods for transmitting data in a wireless communication network

ABSTRACT

A method of communicating over a wideband communication channel divided into a plurality of sub-channels comprises dividing a single serial message intended for one of the plurality of communication devices into a plurality of parallel messages, encoding each of the plurality of parallel messages onto at least some of the plurality of sub-channels, and transmitting the encoded plurality of parallel messages to the communication device over the wideband communication channel.

RELATED APPLICATIONS INFORMATION

This application is claims priority under 35 U.S.C. §120 as acontinuation in part to U.S. patent application Ser. No. 10/120,456,entitled “Systems and Methods for Recovering Bandwidth in a WirelessCommunication Network,” filed Apr. 9, 2002, which is in turn acontinuation in part of U.S. patent application Ser. No. 10/010,601,entitled “Systems and Methods for Wireless Communication Over A WideBandwidth Channel Using A Plurality of Sub-channels,” filed on Dec. 6,2001, now abandoned both of which are incorporated herein by referencein the entirety as if set forth in full.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates generally to wireless communication and moreparticularly to systems and methods for wireless communication over awide bandwidth channel using a plurality of sub-channels.

2. Background

Wireless communication systems are proliferating at the Wide AreaNetwork (WAN), Local Area Network (LAN), and Personal Area Network (PAN)levels. These wireless communication systems use a variety of techniquesto allow simultaneous access to multiple users. The most common of thesetechniques are Frequency Division Multiple Access (FDMA), which assignsspecific frequencies to each user, Time Division Multiple Access (TDMA),which assigns particular time slots to each user, and Code DivisionMultple Access (CDMA), which assigns specific codes to each user. Butthese wireless communication systems and various modulation techniquesare afflicted by a host of problems that limit the capacity and thequality of service provided to the users. The following paragraphsbriefly describe a few of these problems for the purpose ofillustration.

One problem that can exist in a wireless communication system ismultipath interference. Multipath interference, or multipath, occursbecause some of the energy in a transmitted wireless signal bounces offof obstacles, such as buildings or mountains, as it travels from sourceto destination. The obstacles in effect create reflections of thetransmitted signal and the more obstacles there are, the morereflections they generate. The reflections then travel along their owntransmission paths to the destination (or receiver). The reflectionswill contain the same information as the original signal; however,because of the differing transmission path lengths, the reflectedsignals will be out of phase with the original signal. As a result, theywill often combine destructively with the original signal in thereceiver. This is referred to as fading. To combat fading, currentsystems typically try to estimate the multipath effects and thencompensate for them in the receiver using an equalizer. In practice,however, it is very difficult to achieve effective multipathcompensation.

A second problem that can affect the operation of wireless communicationsystems is interference from adjacent communication cells within thesystem. In FDMA/TDMA systems, this type of interference is preventedthrough a frequency reuse plan. Under a frequency reuse plan, availablecommunication frequencies are allocated to communication cells withinthe communication system such that the same frequency will not be usedin adjacent cells. Essentially, the available frequencies are split intogroups. The number of groups is termed the reuse factor. Then thecommunication cells are grouped into clusters, each cluster containingthe same number of cells as there are frequency groups. Each frequencygroup is then assigned to a cell in each cluster. Thus, if a frequencyreuse factor of 7 is used, for example, then a particular communicationfrequency will be used only once in every seven communication cells. Asa result, in any group of seven communication cells, each cell can onlyuse 1/7^(th) of the available frequencies, i.e., each cell is only ableto use 1/7^(th) of the available bandwidth.

In a CDMA communication system, each cell uses the same widebandcommunication channel. In order to avoid interference with adjacentcells, each communication cell uses a particular set of spread spectrumcodes to differentiate communications within the cell from thoseoriginating outside of the cell. Thus, CDMA systems preserve thebandwidth in the sense that they avoid limitations inherent toconventional reuse planning. But as will be discussed, there are otherissues that limit the bandwidth in CDMA systems as well.

Thus, in overcoming interference, system bandwidth is often sacrificed.Bandwidth is becoming a very valuable commodity as wirelesscommunication systems continue to expand by adding more and more users.Therefore, trading off bandwidth for system performance is a costly,albeit necessary, proposition that is inherent in all wirelesscommunication systems.

The foregoing are just two examples of the types of problems that canaffect conventional wireless communication systems. The examples alsoillustrate that there are many aspects of wireless communication systemperformance that can be improved through systems and methods that, forexample, reduce interference, increase bandwidth, or both.

Not only are conventional wireless communication systems effected byproblems, such as those described in the preceding paragraphs, but alsodifferent types of systems are effected in different ways and todifferent degrees. Wireless communication systems can be split intothree types: 1) line-of-sight systems, which can include point-to-pointor point-to-multipoint systems; 2) indoor non-line of sight systems; and3) outdoor systems such as wireless WANs. Line-of-sight systems areleast affected by the problems described above, while indoor systems aremore affected, due for example to signals bouncing off of buildingwalls. Outdoor systems are by far the most affected of the threesystems. Because these types of problems are limiting factors in thedesign of wireless transmitters and receivers, such designs must betailored to the specific types of system in which it will operate. Inpractice, each type of system implements unique communication standardsthat address the issues unique to the particular type of system. Even ifan indoor system used the same communication protocols and modulationtechniques as an outdoor system, for example, the receiver designs wouldstill be different because multipath and other problems are unique to agiven type of system and must be addressed with unique solutions. Thiswould not necessarily be the case if cost efficient and effectivemethodologies can be developed to combat such problems as describedabove that build in programmability so that a device can be reconfiguredfor different types of systems and still maintain superior performance.

SUMMARY OF THE INVENTION

In order to combat the above problems, the systems and methods describedherein provide a novel channel access technology that provides a costefficient and effective methodology that builds in programmability sothat a device can be reconfigured for different types of systems andstill maintain superior performance. In one aspect of the invention, amethod of communicating over a wideband-communication channel dividedinto a plurality of sub-channels is provided. The method comprisesdividing a single serial message intended for one of the plurality ofcommunication devices into a plurality of parallel messages, encodingeach of the plurality of parallel messages onto at least some of theplurality of sub-channels, and transmitting the encoded plurality ofparallel messages to the communication device over the widebandcommunication channel.

When symbols are restricted to particular range of values, thetransmitters and receivers can be simplified to eliminate high powerconsuming components such as a local oscillator, synthesizer and phaselocked loops. Thus, in one aspect a transmitter comprises a plurality ofpulse converters and differential amplifiers, to convert a balancedtrinary data stream into a pulse sequence which can be filtered toreside in the desired frequency ranges and phase. The use of thebalanced trinary data stream allows conventional components to bereplaced by less costly, smaller components that consume less power.

Simialrly, in another aspect, a receiver comprises detection of themagnitude and phase of the symbols, which can be achieved with anenvelope detector and sign detector respectively. Thus, conventionalreceiver components can be replaced by less costly, smaller componentsthat consume less power.

Other aspects, advantages, and novel features of the invention willbecome apparent from the following Detailed Description of PreferredEmbodiments, when considered in conjunction with the accompanyingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

Preferred embodiments of the present inventions taught herein areillustrated by way of example, and not by way of limitation, in thefigures of the accompanying drawings, in which:

FIG. 1 is a diagram illustrating an example embodiment of a widebandchannel divided into a plurality of sub-channels in accordance with theinvention;

FIG. 2 is a diagram illustrating the effects of multipath in a wirelesscommunication system;

FIG. 3 is a diagram illustrating another example embodiment of awideband communication channel divided into a plurality of sub-channelsin accordance with the invention;

FIG. 4 is a diagram illustrating the application of a roll-off factor tothe sub-channels of FIGS. 1, 2 and 3;

FIG. 5A is a diagram illustrating the assignment of sub-channels for awideband communication channel in accordance with the invention;

FIG. 5B is a diagram illustrating the assignment of time slots for awideband communication channel in accordance with the invention;

FIG. 6 is a diagram illustrating an example embodiment of a wirelesscommunication in accordance with the invention;

FIG. 7 is a diagram illustrating the use of synchronization codes in thewireless communication system of FIG. 6 in accordance with theinvention;

FIG. 8 is a diagram illustrating a correlator that can be used tocorrelate synchronization codes in the wireless communication system ofFIG. 6;

FIG. 9 is a diagram illustrating synchronization code correlation inaccordance with the invention;

FIG. 10 is a diagram illustrating the cross-correlation properties ofsynchronization codes configured in accordance with the invention;

FIG. 11 is a diagram illustrating another example embodiment of awireless communication system in accordance with the invention;

FIG. 12A is a diagram illustrating how sub-channels of a widebandcommunication channel according to the present invention can be groupedin accordance with the present invention;

FIG. 12B is a diagram illustrating the assignment of the groups ofsub-channels of FIG. 12A in accordance with the invention;

FIG. 13 is a diagram illustrating the group assignments of FIG. 12B inthe time domain;

FIG. 14 is a flow chart illustrating the assignment of sub-channelsbased on SIR measurements in the wireless communication system of FIG.11 in accordance with the invention;

FIG. 15 is a logical block diagram of an example embodiment oftransmitter configured in accordance with the invention;

FIG. 16 is a logical block diagram of an example embodiment of amodulator configured in accordance with the present invention for use inthe transmitter of FIG. 15;

FIG. 17 is a diagram illustrating an example embodiment of a ratecontroller configured in accordance with the invention for use in themodulator of FIG. 16;

FIG. 18 is a diagram illustrating another example embodiment of a ratecontroller configured in accordance with the invention for use in themodulator of FIG. 16;

FIG. 19 is a diagram illustrating an example embodiment of a frequencyencoder configured in accordance with the invention for use in themodulator of FIG. 16;

FIG. 20 is a logical block diagram of an example embodiment of a TDM/FDMblock configured in accordance with the invention for use in themodulator of FIG. 16;

FIG. 21 is a logical block diagram of another example embodiment of aTDM/FDM block configured in accordance with the invention for use in themodulator of FIG. 16;

FIG. 22 is a logical block diagram of an example embodiment of afrequency shifter configured in accordance with the invention for use inthe modulator of FIG. 16;

FIG. 23 is a logical block diagram of a receiver configured inaccordance with the invention;

FIG. 24 is a logical block diagram of an example embodiment of ademodulator configured in accordance with the invention for use in thereceiver of FIG. 23;

FIG. 25 is a logical block diagram of an example embodiment of anequalizer configured in accordance with the present invention for use inthe demodulator of FIG. 24;

FIG. 26 is a logical block diagram of an example embodiment of awireless communication device configured in accordance with theinvention;

FIG. 27 is a flow chart illustrating an exemplary method for recoveringbandwidth in a wireless communication network in accordance with theinvention;

FIG. 28 is a diagram illustrating an exemplary wireless communicationnetwork in which the method of FIG. 27 can be implemented;

FIG. 29 is a logical block diagram illustrating an exemplary transmitterthat can be used in the network of FIG. 28 to implement the method ofFIG. 27;

FIG. 30 is a logical block diagram illustrating another exemplarytransmitter that can be used in the network of FIG. 28 to implement themethod of FIG. 27;

FIG. 31 is a diagram illustrating another exemplary wirelesscommunication network in which the method of FIG. 27 can be implemented;

FIG. 32 is a diagram illustrating an exemplary radio transmitter modulethat can be used in the transmitter of FIG. 15;

FIG. 33 is a diagram of an example radio transmitter module that can beused in the transmitter of FIG. 15 in accordance with one embodiment ofthe invention;

FIG. 34 is a waveform illustrating encoding of data bits in the radiotransmitter module of FIG. 33;

FIG. 35 is a diagram illustrating one example embodiment of a pulserthat can be used in the radio transmitter module of FIG. 33;

FIG. 36 is a diagram illustrating an example embodiment of a passivecombiner that can be used in the radio transmitter module of FIG. 33;

FIG. 37 is a diagram illustrating another example embodiment of a pulserthat can be used in the radio transmitter module of FIG. 33;

FIG. 38 is a diagram illustrating an example embodiment of a passivecombiner that can be used in the radio transmitter module of FIG. 33;

FIG. 39 is a diagram illustrating an exemplary radio receiver that canbe used in the receiver of FIG. 23;

FIG. 40 is a diagram of an example radio receiver that can be used inthe receiver of FIG. 23 in accordance with one embodiment of theinvention;

FIG. 41 is a diagram illustrating another example embodiment of areceiver that can be used in the receiver of FIG. 23 in accordance withone embodiment of the invention; and

FIG. 42 is a table illustrating the effective number of bits for variousimplementations of the radio receiver of FIG. 41.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

1. Introduction

In order to improve wireless communication system performance and allowa single device to move from one type of system to another, while stillmaintaining superior performance, the systems and methods describedherein provide various communication methodologies that enhanceperformance of transmitters and receivers with regard to various commonproblems that afflict such systems and that allow the transmittersand/or receivers to be reconfigured for optimal performance in a varietyof systems. Accordingly, the systems and methods described herein definea channel access protocol that uses a common wideband communicationchannel for all communication cells. The wideband channel, however, isthen divided into a plurality of sub-channels. Different sub-channelsare then assigned to one or more users within each cell. But the basestation, or service access point, within each cell transmits one messagethat occupies the entire bandwidth of the wideband channel. Each user'scommunication device receives the entire message, but only decodes thoseportions of the message that reside in sub-channels assigned to theuser. For a point-to-point system, for example, a single user may beassigned all sub-channels and, therefore, has the full wide band channelavailable to them. In a wireless WAN, on the other hand, thesub-channels may be divided among a plurality of users.

In the descriptions of example embodiments that follow, implementationdifferences, or unique concerns, relating to different types of systemswill be pointed out to the extent possible. But it should be understoodthat the systems and methods described herein are applicable to any typeof communication systems. In addition, terms such as communication cell,base station, service access point, etc. are used interchangeably torefer to the common aspects of networks at these different levels.

To begin illustrating the advantages of the systems and methodsdescribed herein, one can start by looking at the multipath effects fora single wideband communication channel 100 of bandwidth B as shown inFIG. 1. Communications sent over channel 100 in a traditional wirelesscommunication system will comprise digital data symbols, or symbols,that are encoded and modulated onto a RF carrier that is centered atfrequency f_(c) and occupies bandwidth B. Generally, the width of thesymbols (or the symbol duration) T is defined as 1/B. Thus, if thebandwidth B is equal to 100 MHz, then the symbol duration T is definedby the following equation:T=1/B=1/100MHZ=10ns.  (1)

When a receiver receives the communication, demodulates it, and thendecodes it, it will recreate a stream 104 of data symbols 106 asillustrated in FIG. 2. But the receiver will also receive multipathversions 108 of the same data stream. Because multipath data streams 108are delayed in time relative to data stream 104 by delays d1, d2, d3,and d4, for example, they may combine destructively with data stream104.

A delay spread d_(s) is defined as the delay from reception of datastream 104 to the reception of the last multipath data stream 108 thatinterferes with the reception of data stream 104. Thus, in the exampleillustrated in FIG. 2, the delay spread d_(s) is equal to delay d4. Thedelay spread d_(s) will vary for different environments. An environmentwith a lot of obstacles will create a lot of multipath reflections.Thus, the delay spread d_(s) will be longer. Experiments have shown thatfor outdoor WAN type environments, the delay spread d_(s) can be as longas 20 μs. Using the 10 ns symbol duration of equation (1), thistranslates to 2000 symbols. Thus, with a very large bandwidth, such as100 MHz, multipath interference can cause a significant amount ofinterference at the symbol level for which adequate compensation isdifficult to achieve. This is true even for indoor environments. Forindoor LAN type systems, the delay spread d_(s) is significantlyshorter, typically about 1 μs. For a 10 ns symbol duration, this isequivalent to 100 symbols, which is more manageable but stillsignificant.

By segmenting the bandwidth B into a plurality of sub-channels 200, asillustrated in FIG. 3, and generating a distinct data stream for eachsub-channel, the multipath effect can be reduced to a much moremanageable level. For example, if the bandwidth B of each sub-channel200 is 500 KHz, then the symbol duration is 2 μs. Thus, the delay spreadd_(s) for each sub-channel is equivalent to only 10 symbols (outdoor) orhalf a symbol (indoor). Thus, by breaking up a message that occupies theentire bandwidth B into discrete messages, each occupying the bandwidthB of sub-channels 200, a very wideband signal that suffers fromrelatively minor multipath effects is created.

Before discussing further features and advantages of using a widebandcommunication channel segmented into a plurality of sub-channels asdescribed, certain aspects of the sub-channels will be explained in moredetail. Referring back to FIG. 3, the overall bandwidth B is segmentedinto N sub-channels center at frequencies f₀ to f_(N−1). Thus, thesub-channel 200 that is immediately to the right of fc is offset from fcby b/2, where b is the bandwidth of each sub-channel 200. The nextsub-channel 200 is offset by 3b/2, the next by 5b/2, and so on. To theleft of fc, each sub-channel 200 is offset by −b/s, −3b/s, −5b/2, etc.

Preferably, sub-channels 200 are non-overlapping as this allows eachsub-channel to be processed independently in the receiver. To accomplishthis, a roll-off factor is preferably applied to the signals in eachsub-channel in a pulse-shaping step. The effect of such a pulse-shapingstep is illustrated in FIG. 3 by the non-rectangular shape of the pulsesin each sub-channel 200. Thus, the bandwidth B of each sub-channel canbe represented by an equation such as the following:b=(1+r)/T;  (2)

Where r=the roll-off factor; and

-   -   T=the symbol duration.

Without the roll-off factor, i.e., b=1T, the pulse shape would berectangular in the frequency domain, which corresponds to a (sin x)/xfunction in the time domain. The time domain signal for a (sin x)/xsignal 400 is shown in FIG. 4 in order to illustrate the problemsassociated with a rectangular pulse shape and the need to use a roll-offfactor.

As can be seen, main lobe 402 comprises almost all of signal 400. Butsome of the signal also resides in side lobes 404, which stretch outindefinitely in both directions from main lobe 402. Side lobes 404 makeprocessing signal 400 much more difficult, which increases thecomplexity of the receiver. Applying a roll-off factor r, as in equation(2), causes signal 400 to decay faster, reducing the number of sidelobes 404. Thus, increasing the roll-off factor decreases the length ofsignal 400, i.e., signal 400 becomes shorter in time. But including theroll-off factor also decreases the available bandwidth in eachsub-channel 200. Therefore, r must be selected so as to reduce thenumber of side lobes 404 to a sufficient number, e.g., 15, while stillmaximizing the available bandwidth in each sub-channel 200.

Thus, the overall bandwidth B for communication channel 200 is given bythe following equation:B=N(1+r)/T;  (3)

orB=M/T;  (4)

WhereM=(1+r)N.  (5)

For efficiency purposes related to transmitter design, it is preferablethat r is chosen so that M in equation (5) is an integer. Choosing r sothat M is an integer allows for more efficient transmitters designsusing, for example, Inverse Fast Fourier Transform (IFFT) techniques.Since M=N+N(r), and N is always an integer, this means that r must bechosen so that N(r) is an integer. Generally, it is preferable for r tobe between 0.1 and 0.5. Therefore, if N is 16, for example, then 0.5could be selected for r so that N(r) is an integer. Alternatively, if avalue for r is chosen in the above example so that N(r) is not aninteger, B can be made slightly wider than MIT to compensate. In thiscase, it is still preferable that r be chosen so that N(r) isapproximately an integer.

2. Example Embodiment of a Wireless Communication System

With the above in mind, FIG. 6 illustrates an example communicationsystem 600 comprising a plurality of cells 602 that each use a commonwideband communication channel to communicate with communication devices604 within each cell 602. The common communication channel is a widebandcommunication channel as described above. Each communication cell 602 isdefined as the coverage area of a base station, or service access point,606 within the cell. One such base station 606 is shown for illustrationin FIG. 6. For purposes of this specification and the claims thatfollow, the term base station will be used generically to refer to adevice that provides wireless access to the wireless communicationsystem for a plurality of communication devices, whether the system is aline of sight, indoor, or outdoor system.

Because each cell 602 uses the same communication channel, signals inone cell 602 must be distinguishable from signals in adjacent cells 602.To differentiate signals from one cell 602 to another, adjacent basestations 606 use different synchronization codes according to a codereuse plan. In FIG. 6, system 600 uses a synchronization code reusefactor of 4, although the reuse factor can vary depending on theapplication.

Preferably, the synchronization code is periodically inserted into acommunication from a base station 606 to a communication device 604 asillustrated in FIG. 7. After a predetermined number of data packets 702,in this case two, the particular synchronization code 704 is insertedinto the information being transmitted by each base station 606. Asynchronization code is a sequence of data bits known to both the basestation 606 and any communication devices 604 with which it iscommunicating. The synchronization code allows such a communicationdevice 604 to synchronize its timing to that of base station 606, which,in turn, allows device 604 to decode the data properly. Thus, in cell 1(see lightly shaded cells 602 in FIG. 6), for example, synchronizationcode 1 (SYNC1) is inserted into data stream 706, which is generated bybase station 606 in cell 1, after every two packets 702; in cell 2 SYNC2is inserted after every two packets 702; in cell 3 SYNC3 is inserted;and in cell 4 SYNC4 is inserted. Use of the synchronization codes isdiscussed in more detail below.

In FIG. 5A, an example wideband communication channel 500 for use incommunication system 600 is divided into 16 sub-channels 502, centeredat frequencies f₀ to f₁₅. A base station 606 at the center of eachcommunication cell 602 transmits a single packet occupying the wholebandwidth B of wideband channel 500. Such a packet is illustrated bypacket 504 in FIG. 5B. Packet 504 comprises sub-packets 506 that areencoded with a frequency offset corresponding to one of sub-channels502. Sub-packets 506 in effect define available time slots in packet504. Similarly, sub-channels 502 can be said to define availablefrequency bins in communication channel 500. Therefore, the resourcesavailable in communication cell 602 are time slots 506 and frequencybins 502, which can be assigned to different communication devices 604within each cell 602.

Thus, for example, frequency bins 502 and time slots 506 can be assignedto 4 different communication devices 604 within a cell 602 as shown inFIG. 5B. Each communication device 604 receives the entire packet 504,but only processes those frequency bins 502 and/or timeslots 506 thatare assigned to it. Preferably, each device 604 is assigned non-adjacentfrequency bins 502, as in FIG. 5. This way, if interference corrupts theinformation in a portion of communication channel 500, then the effectsare spread across all devices 604 within a cell 602. Hopefully, byspreading out the effects of interference in this manner the effects areminimized and the entire information sent to each device 604 can stillbe recreated from the unaffected information received in other frequencybins. For example, if interference, such as fading, corrupted theinformation in bins f₀-f₄ then each user 1-4 loses one packet of data.But each user potentially receives three unaffected packets from theother bins assigned to them. Hopefully, the unaffected data in the otherthree bins provides enough information to recreate the entire messagefor each user. Thus, frequency diversity can be achieved by assigningnon-adjacent bins to each of multiple users.

Ensuring that the bins assigned to one user are separated by more thanthe coherence bandwidth ensures frequency diversity. As discussed above,the coherence bandwidth is approximately equal to 1/d_(s). For outdoorsystems, where ds is typically 1 μs, 1/d_(s)=1/1 μs=1 MHz. Thus, thenon-adjacent frequency bands assigned to a user are preferably separatedby at least 1 MHz. It can be even more preferable, however, if thecoherence bandwidth plus some guard band to ensure sufficient frequencydiversity separate the non-adjacent bins assigned to each user. Forexample, it is preferable in certain implementations to ensure that atleast 5 times the coherence bandwidth, or 5 MHz in the above example,separates the non-adjacent bins.

Another way to provide frequency diversity is to repeat blocks of datain frequency bins assigned to a particular user that are separated bymore than the coherence bandwidth. In other words, if 4 sub-channels 200are assigned to a user, then data block a can be repeated in the firstand third sub-channels 200 and data block b can be repeated in thesecond and fourth sub-channels 202, provided the sub-channels aresufficiently separated in frequency. In this case, the system can besaid to be using a diversity length factor of 2. The system cansimilarly be configured to implement other diversity lengths, e.g., 3,4, . . . , l.

It should be noted that spatial diversity can also be included dependingon the embodiment. Spatial diversity can comprise transmit spatialdiversity, receive spatial diversity, or both. In transmit spatialdiversity, the transmitter uses a plurality of separate transmitters anda plurality of separate antennas to transmit each message. In otherwords, each transmitter transmits the same message in parallel. Themessages are then received from the transmitters and combined in thereceiver. Because the parallel transmissions travel different paths, ifone is affected by fading, the others will likely not be affected. Thus,when they are combined in the receiver, the message should berecoverable even if one or more of the other transmission pathsexperienced severe fading.

Receive spatial diversity uses a plurality of separate receivers and aplurality of separate antennas to receive a single message. If anadequate distance separates the antennas, then the transmission path forthe signals received by the antennas will be different. Again, thisdifference in the transmission paths will provide imperviousness tofading when the signals from the receivers are combined.

Transmit and receive spatial diversity can also be combined within asystem such as system 600 so that two antennas are used to transmit andtwo antennas are used to receive. Thus, each base station 606transmitter can include two antennas, for transmit spatial diversity,and each communication device 604 receiver can include two antennas, forreceive spatial diversity. If only transmit spatial diversity isimplemented in system 600, then it can be implemented in base stations606 or in communication devices 604. Similarly, if only receive spatialdiversity is included in system 600, then it can be implemented in basestations 606 or communication devices 604.

The number of communication devices 604 assigned frequency bins 502and/or time slots 506 in each cell 602 is preferably programmable inreal time. In other words, the resource allocation within acommunication cell 602 is preferably programmable in the face of varyingexternal conditions, i.e., multipath or adjacent cell interference, andvarying requirements, i.e., bandwidth requirements for various userswithin the cell. Thus, if user 1 requires the whole bandwidth todownload a large video file, for example, then the allocation of bins502 can be adjust to provide user 1 with more, or even all, of bins 502.Once user 1 no longer requires such large amounts of bandwidth, theallocation of bins 502 can be readjusted among all of users 1-4.

It should also be noted that all of the bins assigned to a particularuser can be used for both the forward and reverse link. Alternatively,some bins 502 can be assigned as the forward link and some can beassigned for use on the reverse link, depending on the implementation.

To increase capacity, the entire bandwidth B is preferably reused ineach communication cell 602, with each cell 602 being differentiated bya unique synchronization code (see discussion below). Thus, system 600provides increased immunity to multipath and fading as well as increasedband width due to the elimination of frequency reuse requirements.

3. Synchronization

FIG. 8 illustrates an example embodiment of a synchronization codecorrelator 800. When a device 604 in cell 1 (see FIG. 6), for example,receives an incoming communication from the cell 1 base station 606, itcompares the incoming data with SYNC1 in correlator 800. Essentially,the device scans the incoming data trying to correlate the data with theknown synchronization code, in this case SYNC1. Once correlator 800matches the incoming data to SYNC1 it generates a correlation peak 804at the output. Multipath versions of the data will also generatecorrelation peaks 806, although these peaks 806 are generally smallerthan correlation peak 804. The device can then use the correlation peaksto perform channel estimation, which allows the device to adjust for themultipath using, e.g., an equalizer. Thus, in cell 1, if correlator 800receives a data stream comprising SYNC1, it will generate correlationpeaks 804 and 806. If, on the other hand, the data stream comprisesSYNC2, for example, then no peaks will be generated and the device willessentially ignore the incoming communication.

Even though a data stream that comprises SYNC2 will not create anycorrelation peaks, it can create noise in correlator 800 that canprevent detection of correlation peaks 804 and 806. Several steps can betaken to prevent this from occurring. One way to minimize the noisecreated in correlator 800 by signals from adjacent cells 602, is toconfigure system 600 so that each base station 606 transmits at the sametime. This way, the synchronization codes can preferably be generated insuch a manner that only the synchronization codes 704 of adjacent celldata streams, e.g., streams 708, 710, and 712, as opposed to packets 702within those streams, will interfere with detection of the correctsynchronization code 704, e.g., SYNC1. The synchronization codes canthen be further configured to eliminate or reduce the interference.

For example, the noise or interference caused by an incorrectsynchronization code is a function of the cross correlation of thatsynchronization code with respect to the correct code. The better thecross correlation between the two, the lower the noise level. When thecross correlation is ideal, then the noise level will be virtually zeroas illustrated in FIG. 9 by noise level 902. Therefore, a preferredembodiment of system 600 uses synchronization codes that exhibit idealcross correlation, i.e., zero. Preferably, the ideal cross correlationof the synchronization codes covers a period 1 that is sufficient toallow accurate detection of multipath correlation peaks 906 as well ascorrelation peak 904. This is important so that accurate channelestimation and equalization can take place. Outside of period 1, thenoise level 908 goes up, because the data in packets 702 is random andwill exhibit low cross correlation with the synchronization code, e.g.,SYNC1. Preferably, period 1 is actually slightly longer then themultipath length in order to ensure that the multipath can be detected.

a. Synchronization Code Generation

Conventional systems use orthogonal codes to achieve cross correlationin correlator 800. In system 600 for example, SYNC1, SYNC2, SYNC3, andSYNC4, corresponding to cells 1-4 (see lightly shaded cells 602 of FIG.6) respectively, will all need to be generated in such a manner thatthey will have ideal cross correlation with each other. In oneembodiment, if the data streams involved comprise high and low databits, then the value “1” can be assigned to the high data bits and “−1”to the low data bits. Orthogonal data sequences are then those thatproduce a “0” output when they are exclusively ORed (XORed) together incorrelator 800. The following example illustrates this point fororthogonal sequences 1 and 2:

$\;{\frac{\begin{matrix}{{sequence}\mspace{14mu} 1\text{:}} & 1 & 1 & {- 1} & {\mspace{14mu} 1} \\{{sequence}\mspace{14mu} 2\text{:}} & 1 & 1 & {\mspace{14mu} 1} & {- 1}\end{matrix}}{\mspace{146mu}\begin{matrix}1 & 1 & {- 1} & {- 1}\end{matrix}} = 0}$

Thus, when the results of XORing each bit pair are added, the result is“0.”

But in system 600, for example, each code must have ideal, or zero,cross correlation with each of the other codes used in adjacent cells602. Therefore, in one example embodiment of a method for generatingsynchronization codes exhibiting the properties described above, theprocess begins by selecting a “perfect sequence” to be used as the basisfor the codes. A perfect sequence is one that when correlated withitself produces a number equal to the number of bits in the sequence.For example:

${{{Perfect}\mspace{20mu}{sequence}\mspace{20mu} 1}:\frac{\begin{matrix}{\mspace{14mu} 1} & 1 & {- 1} & 1 \\{\mspace{14mu} 1} & 1 & {- 1} & 1\end{matrix}}{\mspace{14mu}\begin{matrix}1 & 1 & {\mspace{14mu} 1} & 1\end{matrix}}} = 4$

But each time a perfect sequence is cyclically shifted by one bit, thenew sequence is orthogonal with the original sequence. Thus, forexample, if perfect sequence 1 is cyclically shifted by one bit and thencorrelated with the original, the correlation produces a “0” as in thefollowing example:

${{{Perfect}\mspace{14mu}{sequence}\mspace{14mu} 1}:\frac{\begin{matrix}1 & 1 & {- 1} & {\mspace{20mu} 1} \\1 & 1 & {\mspace{14mu} 1} & {- 1}\end{matrix}}{\begin{matrix}1 & 1 & {- 1} & {- 1}\end{matrix}}} = 0$

If the perfect sequence 1 is again cyclically shifted by one bit, andagain correlated with the original, then it will produce a “0”. Ingeneral, you can cyclically shift a perfect sequence by any number ofbits up to its length and correlate the shifted sequence with theoriginal to obtain a “0”.

Once a perfect sequence of the correct length is selected, the firstsynchronization code is preferably generated in one embodiment byrepeating the sequence 4 times. Thus, if perfect sequence 1 is beingused, then a first synchronization code y would be the following:y=1 1 −1 1 1 1 −1 1 1 1 −1 1 1 1 −1 1.

Or in generic form:y=x(0)x(1)x(2)x(3)x(0)x(1)x(2)x(3)x(0)x(1)x(2)x(3)x(0)x(1)x(2)x(3).

For a sequence of length L:y=x(0)x(1) . . . x(L)x(0)x(1) . . . x(L)x(0)x(1) . . . x(L)x(0)x(1) . .. x(L).

Repeating the perfect sequence allows correlator 800 a betteropportunity to detect the synchronization code and allows generation ofother uncorrelated frequencies as well. Repeating has the effect ofsampling in the frequency domain. This effect is illustrated by thegraphs in FIG. 10. Thus, in trace 1, which corresponds tosynchronization code y, a sample 1002 is generated every fourth samplebin 1000. Each sample bin is separated by 1/(4L×T), where T is thesymbol duration. Thus in the above example, where L=4, each sample binis separated by 1/(16×T) in the frequency domain. Traces 2-4 illustratethe next three synchronization codes. As can be seen, the samples foreach subsequent synchronization code are shifted by one sample binrelative to the samples for the previous sequence. Therefore, none ofsequences interfere with each other.

To generate the subsequent sequences, corresponding to traces 2-4,sequence y must be shifted in frequency. This can be accomplished usingthe following equation:z ^(r)(m)=y(m)*exp(j*2*πr*m/(n*L)),  (5)

-   -   for r=1 to L (# of sequences) and m=0 to 4*L−1 (time); and    -   where: z^(r)(m)=each subsequent sequence,        -   y(m)=the first sequence, and        -   n=the number of times the sequence is repeated.

It will be understood that multiplying by an exp(j2π(r*m/N)) factor,where N is equal to the number of times the sequence is repeated (n)multiplied by the length of the underlying perfect sequence L, in thetime domain results in a shift in the frequency domain. Equation (5)results in the desired shift as illustrated in FIG. 10 for each ofsynchronization codes 2-4, relative to synchronization code 1. The finalstep in generating each synchronization code is to append the copies ofthe last M samples, where M is the length of the multipath, to the frontof each code. This is done to make the convolution with the multipathcyclic and to allow easier detection of the multipath.

It should be noted that synchronization codes can be generated from morethan one perfect sequence using the same methodology. For example, aperfect sequence can be generated and repeated for times and then asecond perfect sequence can be generated and repeated four times to geta n factor equal to eight. The resulting sequence can then be shifted asdescribed above to create the synchronization codes.

b. Signal Measurements Using Synchronization Codes

Therefore, when a communication device is at the edge of a cell, it willreceive signals from multiple base stations and, therefore, will bedecoding several synchronization codes at the same time. This can beillustrated with the help of FIG. 11, which illustrates another exampleembodiment of a wireless communication system 1100 comprisingcommunication cells 1102, 1104, and 1106 as well as communication device1108, which is in communication with base station 1110 of cell 1102 butalso receiving communication from base stations 1112 and 1114 of cells1104 and 1106, respectively.

If communications from base station 1110 comprise synchronization codeSYNC1 and communications from base station 1112 and 1114 comprise SYNC2and SYNC3 respectively, then device 1108 will effectively receive thesum of these three synchronization codes. This is because, as explainedabove, base stations 1110, 1112, and 1114 are configured to transmit atthe same time. Also, the synchronization codes arrive at device 1108 atalmost the same time because they are generated in accordance with thedescription above.

Again as described above, the synchronization codes SYNC1, SYNC2, andSYNC3 exhibit ideal cross correlation. Therefore, when device 1108correlates the sum x of codes SYNC1, SYNC2, and SYNC3, the latter twowill not interfere with proper detection of SYNC1 by device 1108.Importantly, the sum x can also be used to determine important signalcharacteristics, because the sum x is equal to the sum of thesynchronization code signal in accordance with the following equation:x=SYNC1+SYNC2+SYNC3.  (6)

Therefore, when SYNC1 is removed, the sum of SYNC2 and SYNC3 is left, asshown in the following:x−SYNC1=SYNC2+SYNC3.  (7)

The energy computed from the sum (SYNC2+SYNC3) is equal to the noise orinterference seen by device 1108. Since the purpose of correlating thesynchronization code in device 1106 is to extract the energy in SYNC1,device 1108 also has the energy in the signal from base station 1110,i.e., the energy represented by SYNC1. Therefore, device 1106 can usethe energy of SYNC1 and of (SYNC2+SYNC3) to perform asignal-to-interference measurement for the communication channel overwhich it is communicating with base station 1110. The result of themeasurement is preferably a signal-to-interference ratio (SIR). The SIRmeasurement can then be communicated back to base station 1110 forpurposes that will be discussed below.

The ideal cross correlation of the synchronization codes also allowsdevice 1108 to perform extremely accurate determinations of the ChannelImpulse Response (CIR), or channel estimation, from the correlationproduced by correlator 800. This allows for highly accurate equalizationusing low cost, low complexity equalizers, thus overcoming a significantdraw back of conventional systems.

4. Sub-channel Assignments

As mentioned, the SIR as determined by device 1108 can be communicatedback to base station 1110 for use in the assignment of slots 502. In oneembodiment, due to the fact that each sub-channel 502 is processedindependently, the SIR for each sub-channel 502 can be measured andcommunicated back to base station 1110. In such an embodiment,therefore, sub-channels 502 can be divided into groups and a SIRmeasurement for each group can be sent to base station 1110. This isillustrated in FIG. 12A, which shows a wideband communication channel1200 segmented into sub-channels f₀ to f₁₅. Sub-channels f₀ to f₁₅ arethen grouped into 8 groups G1 to G8. Thus, in one embodiment, device1108 and base station 1110 communicate over a channel such as channel1200.

Sub-channels in the same group are preferably separated by as manysub-channels as possible to ensure diversity. In FIG. 12A for example,sub-channels within the same group are 7 sub-channels apart, e.g., groupG1 comprises f₀ and f₈.

Device 1102 reports a SIR measurement for each of the groups G1 to G8.These SIR measurements are preferably compared with a threshold value todetermine which sub-channels groups are useable by device 1108. Thiscomparison can occur in device 1108 or base station 1110. If it occursin device 1108, then device 1108 can simply report to base station 1110which sub-channel groups are useable by device 1108.

SIR reporting will be simultaneously occurring for a plurality ofdevices within cell 1102. Thus, FIG. 12B illustrates the situation wheretwo communication devices corresponding to user1 and user2 report SIRlevels above the threshold for groups G1, G3, G5, and G7. Base station1110 preferably then assigns sub-channel groups to user1 and user2 basedon the SIR reporting as illustrated in FIG. 12B. When assigning the“good” sub-channel groups to user1 and user2, base station 1110 alsopreferably assigns them based on the principles of frequency diversity.In FIG. 12B, therefore, user1 and user2 are alternately assigned everyother “good” sub-channel.

The assignment of sub-channels in the frequency domain is equivalent tothe assignment of time slots in the time domain. Therefore, asillustrated in FIG. 13, two users, user1 and user2, receive packet 1302transmitted over communication channel 1200. FIG. 13 also illustratedthe sub-channel assignment of FIG. 12B. While FIG. 12 and 13 illustratesub-channel/time slot assignment based on SIR for two users, theprinciples illustrated can be extended for any number of users. Thus, apacket within cell 1102 can be received by 3 or more users. Although, asthe number of available sub-channels is reduced due to high SIR, so isthe available bandwidth. In other words, as available sub-channels arereduced, the number of users that can gain access to communicationchannel 1200 is also reduced.

Poor SIR can be caused for a variety of reasons, but frequently itresults from a device at the edge of a cell receiving communicationsignals from adjacent cells. Because each cell is using the samebandwidth B, the adjacent cell signals will eventually raise the noiselevel and degrade SIR for certain sub-channels. In certain embodiments,therefore, sub-channel assignment can be coordinated between cells, suchas cells 1102, 1104, and 1106 in FIG. 11, in order to preventinterference from adjacent cells.

Thus, if communication device 1108 is near the edge of cell 1102, anddevice 1118 is near the edge of cell 1106, then the two can interferewith each other. As a result, the SIR measurements that device 1108 and1118 report back to base stations 1110 and 1114, respectively, willindicate that the interference level is too high. Base station 1110 canthen be configured to assign only the odd groups, i.e., G1, G3, G5,etc., to device 1108, while base station 1114 can be configured toassign the even groups to device 1118 in a coordinated fashion. The twodevices 1108 and 1118 will then not interfere with each other due to thecoordinated assignment of sub-channel groups.

Assigning the sub-channels in this manner reduces the overall bandwidthavailable to devices 1108 and 1118, respectively. In this case thebandwidth is reduced by a factor of two. But it should be rememberedthat devices operating closer to each base station 1110 and 1114,respectively, will still be able to use all sub-channels if needed.Thus, it is only devices, such as device 1108, that are near the edge ofa cell that will have the available bandwidth reduced. Contrast thiswith a CDMA system, for example, in which the bandwidth for all users isreduced, due to the spreading techniques used in such systems, byapproximately a factor of 10 at all times. It can be seen, therefore,that the systems and methods for wireless communication over a widebandwidth channel using a plurality of sub-channels not only improvesthe quality of service, but can also increase the available bandwidthsignificantly.

When there are three devices 1108, 1118, and 1116 near the edge of theirrespective adjacent cells 1102, 1104, and 1106, the sub-channels can bedivided by three. Thus, device 1108, for example, can be assigned groupsG1, G4, etc., device 1118 can be assigned groups G2, G5, etc., anddevice 1116 can be assigned groups G3, G6, etc. In this case theavailable bandwidth for these devices, i.e., devices near the edges ofcells 1102, 1104, and 1106, is reduced by a factor of 3, but this isstill better than a CDMA system, for example.

The manner in which such a coordinated assignment of sub-channels canwork is illustrated by the flow chart in FIG. 14. First in step 1402, acommunication device, such as device 1108, reports the SIR for allsub-channel groups G1 to G8. The SIRs reported are then compared, instep 1404, to a threshold to determine if the SIR is sufficiently lowfor each group. Alternatively, device 1108 can make the determinationand simply report which groups are above or below the SIR threshold. Ifthe SIR levels are good for each group, then base station 1110 can makeeach group available to device 1108, in step 1406. Periodically, device1108 preferably measures the SIR level and updates base station 1110 incase the SIR as deteriorated. For example, device 1108 may move fromnear the center of cell 1102 toward the edge, where interference from anadjacent cell may affect the SIR for device 1108.

If the comparison in step 1404 reveals that the SIR levels are not good,then base station 1110 can be preprogrammed to assign either the oddgroups or the even groups only to device 1108, which it will do in step1408. Device 1108 then reports the SIR measurements for the odd or evengroups it is assigned in step 1410, and they are again compared to a SIRthreshold in step 1412.

It is assumed that the poor SIR level is due to the fact that device1108 is operating at the edge of cell 1102 and is therefore beinginterfered with by a device such as device 1118. But device 1108 will beinterfering with device 1118 at the same time. Therefore, the assignmentof odd or even groups in step 1408 preferably corresponds with theassignment of the opposite groups to device 1118, by base station 1114.Accordingly, when device 1108 reports the SIR measurements for whichevergroups, odd or even, are assigned to it, the comparison in step 1410should reveal that the SIR levels are now below the threshold level.Thus, base station 1110 makes the assigned groups available to device1108 in step 1414. Again, device 1108 preferably periodically updatesthe SIR measurements by returning to step 1402.

It is possible for the comparison of step 1410 to reveal that the SIRlevels are still above the threshold, which should indicate that a thirddevice, e.g., device 1116 is still interfering with device 1108. In thiscase, base station 1110 can be preprogrammed to assign every third groupto device 1108 in step 1416. This should correspond with thecorresponding assignments of non-interfering channels to devices 1118and 1116 by base stations 1114 and 1112, respectively. Thus, device 1108should be able to operate on the sub-channel groups assigned, i.e., G1,G4, etc., without undue interference. Again, device 1108 preferablyperiodically updates the SIR measurements by returning to step 1402.Optionally, a third comparison step (not shown) can be implemented afterstep 1416, to ensure that the groups assigned to device 1408 posses anadequate SIR level for proper operation. Moreover, if there are moreadjacent cells, i.e., if it is possible for devices in a 4^(th) or evena 5^(th) adjacent cell to interfere with device 1108, then the processof FIG. 14 would continue and the sub-channel groups would be dividedeven further to ensure adequate SIR levels on the sub-channels assignedto device 1108.

Even though the process of FIG. 14 reduces the bandwidth available todevices at the edge of cells 1102, 1104, and 1106, the SIR measurementscan be used in such a manner as to increase the data rate and thereforerestore or even increase bandwidth. To accomplish this, the transmittersand receivers used in base stations 1102, 1104, and 1106, and in devicesin communication therewith, e.g., devices 1108, 1114, and 1116respectively, must be capable of dynamically changing the symbol mappingschemes used for some or all of the sub-channel. For example, in someembodiments, the symbol mapping scheme can be dynamically changed amongBPSK, QPSK, 8PSK, 16 QAM, 32 QAM, etc. As the symbol mapping schememoves higher, i.e., toward 32 QAM, the SIR level required for properoperation moves higher, i.e., less and less interference can bewithstood. Therefore, once the SIR levels are determined for each group,the base station, e.g., base station 1110, can then determine whatsymbol mapping scheme can be supported for each sub-channel group andcan change the modulation scheme accordingly. Device 1108 must alsochange the symbol mapping scheme to correspond to that of the basestations. The change can be effected for all groups uniformly, or it canbe effected for individual groups. Moreover, the symbol mapping schemecan be changed on just the forward link, just the reverse link, or both,depending on the embodiment.

Thus, by maintaining the capability to dynamically assign sub-channelsand to dynamically change the symbol mapping scheme used for assignedsub-channels, the systems and methods described herein provide theability to maintain higher available bandwidths with higher performancelevels than conventional systems. To fully realize the benefitsdescribed, however, the systems and methods described thus far must becapable of implementation in a cost effect and convenient manner.Moreover, the implementation must include reconfigurability so that asingle device can move between different types of communication systemsand still maintain optimum performance in accordance with the systemsand methods described herein. The following descriptions detail examplehigh level embodiments of hardware implementations configured to operatein accordance with the systems and methods described herein in such amanner as to provide the capability just described above.

5. Sample Transmitter Embodiments

FIG. 15 is logical block diagram illustrating an example embodiment of atransmitter 1500 configured for wireless communication in accordancewith the systems and methods described above. The transmitter could, forexample be within a base station, e.g., base station 606, or within acommunication device, such as device 604. Transmitter 1500 is providedto illustrate logical components that can be included in a transmitterconfigured in accordance with the systems and methods described herein.It is not intended to limit the systems and methods for wirelesscommunication over a wide bandwidth channel using a plurality ofsub-channels to any particular transmitter configuration or anyparticular wireless communication system.

With this in mind, it can be seen that transmitter 1500 comprises aserial-to-parallel converter 1504 configured to receive a serial datastream 1502 comprising a data rate R. Serial-to-parallel converter 1504converts data stream 1502 into N parallel data streams 1504, where N isthe number of sub-channels 200. It should be noted that while thediscussion that follows assumes that a single serial data stream isused, more than one serial data stream can also be used if required ordesired. In any case, the data rate of each parallel data stream 1504 isthen R/N. Each data stream 1504 is then sent to a scrambler, encoder,and interleaver block 1506. Scrambling, encoding, and interleaving arecommon techniques implemented in many wireless communicationtransmitters and help to provide robust, secure communication. Examplesof these techniques will be briefly explained for illustrative purposes.

Scrambling breaks up the data to be transmitted in an effort to smoothout the spectral density of the transmitted data. For example, if thedata comprises a long string of “1”s, there will be a spike in thespectral density. This spike can cause greater interference within thewireless communication system. By breaking up the data, the spectraldensity can be smoothed out to avoid any such peaks. Often, scramblingis achieved by XORing the data with a random sequence.

Encoding, or coding, the parallel bit streams 1504 can, for example,provide Forward Error Correction (FEC). The purpose of FEC is to improvethe capacity of a communication channel by adding some carefullydesigned redundant information to the data being transmitted through thechannel. The process of adding this redundant information is known aschannel coding. Convolutional coding and block coding are the two majorforms of channel coding. Convolutional codes operate on serial data, oneor a few bits at a time. Block codes operate on relatively large(typically, up to a couple of hundred bytes) message blocks. There are avariety of useful convolutional and block codes, and a variety ofalgorithms for decoding the received coded information sequences torecover the original data. For example, convolutional encoding or turbocoding with Viterbi decoding is a FEC technique that is particularlysuited to a channel in which the transmitted signal is corrupted mainlyby additive white gaussian noise (AWGN) or even a channel that simplyexperiences fading.

Convolutional codes are usually described using two parameters: the coderate and the constraint length. The code rate, k/n, is expressed as aratio of the number of bits into the convolutional encoder (k) to thenumber of channel symbols (n) output by the convolutional encoder in agiven encoder cycle. A common code rate is ½, which means that 2 symbolsare produced for every 1-bit input into the coder. The constraint lengthparameter, K, denotes the “length” of the convolutional encoder, i.e.how many k-bit stages are available to feed the combinatorial logic thatproduces the output symbols. Closely related to K is the parameter m,which indicates how many encoder cycles an input bit is retained andused for encoding after it first appears at the input to theconvolutional encoder. The m parameter can be thought of as the memorylength of the encoder.

Interleaving is used to reduce the effects of fading. Interleaving mixesup the order of the data so that if a fade interferes with a portion ofthe transmitted signal, the overall message will not be effected. Thisis because once the message is de-interleaved and decoded in thereceiver, the data lost will comprise non-contiguous portions of theoverall message. In other words, the fade will interfere with acontiguous portion of the interleaved message, but when the message isde-interleaved, the interfered with portion is spread throughout theoverall message. Using techniques such as FEC, the missing informationcan then be filled in, or the impact of the lost data may just benegligible.

After blocks 1506, each parallel data stream 1504 is sent to symbolmappers 1508. Symbol mappers 1508 apply the requisite symbol mapping,e.g., BPSK, QPSK, etc., to each parallel data stream 1504. Symbolmappers 1508 are preferably programmable so that the modulation appliedto parallel data streams can be changed, for example, in response to theSIR reported for each sub-channel 202. It is also preferable, that eachsymbol mapper 1508 be separately programmable so that the optimum symbolmapping scheme for each sub-channel can be selected and applied to eachparallel data stream 1504.

After symbol mappers 1508, parallel data streams 1504 are sent tomodulators 1510. Important aspects and features of example embodimentsof modulators 1510 are described below. After modulators 1510, paralleldata streams 1504 are sent to summer 1512, which is configured to sumthe parallel data streams and thereby generate a single serial datastream 1518 comprising each of the individually processed parallel datastreams 1504. Serial data stream 1518 is then sent to radio module 1512,where it is modulated with an RF carrier, amplified, and transmitted viaantenna 1516 according to known techniques. Radio module embodimentsthat can be used in conjunction with the systems and methods describedherein are described below.

The transmitted signal occupies the entire bandwidth B of communicationchannel 100 and comprises each of the discrete parallel data streams1504 encoded onto their respective sub-channels 102 within bandwidth B.Encoding parallel data streams 1504 onto the appropriate sub-channels102 requires that each parallel data stream 1504 be shifted in frequencyby an appropriate offset. This is achieved in modulator 1510.

FIG. 16 is a logical block diagram of an example embodiment of amodulator 1600 in accordance with the systems and methods describedherein. Importantly, modulator 1600 takes parallel data streams 1602performs Time Division Modulation (TDM) or Frequency Division Modulation(FDM) on each data stream 1602, filters them using filters 1612, andthen shifts each data stream in frequency using frequency shifter 1614so that they occupy the appropriate sub-channel. Filters 1612 apply therequired pulse shapping, i.e., they apply the roll-off factor describedin section 1. The frequency shifted parallel data streams 1602 are thensummed and transmitted. Modulator 1600 can also include rate controller1604, frequency encoder 1606, and interpolators 1610. All of thecomponents shown in FIG. 16 are described in more detail in thefollowing paragraphs and in conjunction with FIGS. 17-23.

FIG. 17 illustrates one example embodiment of a rate controller 1700 inaccordance with the systems and methods described herein. Rate control1700 is used to control the data rate of each parallel data stream 1602.In rate controller 1700, the data rate is halved by repeating datastreams d(0) to d(7), for example, producing streams a(0) to a(15) inwhich a(0) is the same as a(8), a(1) is the same as a(9), etc. FIG. 17illustrates that the effect of repeating the data streams in this manneris to take the data streams that are encoded onto the first 8sub-channels 1702, and duplicate them on the next 8 sub-channels 1702.As can be seen, 7 sub-channels separate sub-channels 1702 comprising thesame, or duplicate, data streams. Thus, if fading effects onesub-channel 1702, for example, the other sub-channels 1702 carrying thesame data will likely not be effected, i.e., there is frequencydiversity between the duplicate data streams. So by sacrificing datarate, in this case half the data rate, more robust transmission isachieved. Moreover, the robustness provided by duplicating the datastreams d(0) to d(8) can be further enhanced by applying scrambling tothe duplicated data streams via scramblers 1704.

It should be noted that the data rate can be reduced by more than half,e.g., by four or more. Alternatively, the data rate can also be reducedby an amount other than half. For example if information from n datastream is encoded onto m sub-channels, where m>n. Thus, to decrease therate by ⅔, information from one data stream can be encoded on a firstsub-channel, information from a second data stream can be encoded on asecond data channel, and the sum or difference of the two data streamscan be encoded on a third channel. In which case, proper scaling willneed to be applied to the power in the third channel. Otherwise, forexample, the power in the third channel can be twice the power in thefirst two.

Preferably, rate controller 1700 is programmable so that the data ratecan be changed responsive to certain operational factors. For example,if the SIR reported for sub-channels 1702 is low, then rate controller1700 can be programmed to provide more robust transmission viarepetition to ensure that no data is lost due to interference.Additionally, different types of wireless communication system, e.g.,indoor, outdoor, line-of-sight, may require varying degrees ofrobustness. Thus, rate controller 1700 can be adjusted to provide theminimum required robustness for the particular type of communicationsystem. This type of programmability not only ensures robustcommunication, it can also be used to allow a single device to movebetween communication systems and maintain superior performance.

FIG. 18 illustrates an alternative example embodiment of a ratecontroller 1800 in accordance with the systems and methods described. Inrate controller 1800 the data rate is increased instead of decreased.This is accomplished using serial-to-parallel converters 1802 to converteach data streams d(0) to d(15), for example, into two data streams.Delay circuits 1804 then delay one of the two data streams generated byeach serial-to-parallel converter 1802 by ½ a symbol, period. Thus, datastreams d(0) to d(15) are transformed into data streams a(0) to a(31).The data streams generated by a particular serial-to-parallel converter1802 and associate delay circuit 1804 must then be summed and encodedonto the appropriate sub-channel. For example, data streams a(0) anda(1) must be summed and encoded onto the first sub-channel. Preferably,the data streams are summed subsequent to each data stream being pulsedshaped by a filter 1612.

Thus, rate controller 1604 is preferably programmable so that the datarate can be increased, as in rate controller 1800, or decreased, as inrate controller 1700, as required by a particular type of wirelesscommunication system, or as required by the communication channelconditions or sub-channel conditions. In the event that the data rate isincreased, filters 1612 are also preferably programmable so that theycan be configured to apply pulse shapping to data streams a(0) to a(31),for example, and then sum the appropriate streams to generate theappropriate number of parallel data streams to send to frequency shifter1614.

The advantage of increasing the data rate in the manner illustrated inFIG. 18 is that higher symbol mapping rates can essentially be achieved,without changing the symbol mapping used in symbol mappers 1508. Oncethe data streams are summed, the summed streams are shifted in frequencyso that they reside in the appropriate sub-channel. But because thenumber of bits per each symbol has been doubled, the symbol mapping ratehas been doubled. Thus, for example, a 4 QAM symbol mapping can beconverted to a 16 QAM symbol mapping, even if the SIR is too high for 16QAM symbol mapping to otherwise be applied. In other words, programmingrate controller 1800 to increase the data rate in the manner illustratedin FIG. 18 can increase the symbol mapping even when channel conditionswould otherwise not allow it, which in turn can allow a communicationdevice to maintain adequate or even superior performance regardless ofthe type of communication system.

The draw back to increasing the data rate as illustrated in FIG. 18 isthat interference is increased, as is receiver complexity. The former isdue to the increased amount of data. The latter is due to the fact thateach symbol cannot be processed independently because of the ½ symboloverlap. Thus, these concerns must be balanced against the increasesymbol mapping ability when implementing a rate controller such as ratecontroller 1800.

FIG. 19 illustrates one example embodiment of a frequency encoder 1900in accordance with the systems and methods described herein. Similar torate encoding, frequency encoding is preferably used to provideincreased communication robustness. In frequency encoder 1900 the sum ordifference of multiple data streams are encoded onto each sub-channel.This is accomplished using adders 1902 to sum data streams d(0) to d(7)with data streams d(8) to d(15), respectively, while adders 1904subtract data streams d(0) to d(7) from data streams d(8) to d(15),respectively, as shown. Thus, data streams a(0) to a(15) generated byadders 1902 and 1904 comprise information related to more than one datastreams d(0) to d(15). For example, a(0) comprises the sum of d(0) andd(8), i.e., d(0)+d(8), while a(8) comprises d(8)−d(0). Therefore, ifeither a(0) or a(8) is not received due to fading, for example, thenboth of data streams d(0) and d(8) can still be retrieved from datastream a(8).

Essentially, the relationship between data stream d(0) to d(15) and a(0)to a(15) is a matrix relationship. Thus, if the receiver knows thecorrect matrix to apply, it can recover the sums and differences of d(0)to d(15) from a(0) to a(15). Preferably, frequency encoder 1900 isprogrammable, so that it can be enabled and disabled in order toprovided robustness when required. Preferable, adders 1902 and 1904 areprogrammable also so that different matrices can be applied to d(0) tod(15).

After frequency encoding, if it is included, data streams 1602 are sentto TDM/FDM blocks 1608. TDM/FDM blocks 1608 perform TDM or FDM on thedata streams as required by the particular embodiment. FIG. 20illustrates an example embodiment of a TDM/FDM block 2000 configured toperform TDM on a data stream. TDM/FDM block 2000 is provided toillustrate the logical components that can be included in a TDM/FDMblock configured to perform TDM on a data stream. Depending on theactual implementation, some of the logical components may or may not beincluded. TDM/FDM block 2000 comprises a sub-block repeater 2002, asub-block scrambler 2004, a sub-block terminator 2006, a sub-blockrepeater 2008, and a SYNC inserter 2010.

Sub-block repeater 2002 is configured to receive a sub-block of data,such as block 2012 comprising bits a(0) to a(3) for example. Sub-blockrepeater is then configured to repeat block 2012 to provide repetition,which in turn leads to more robust communication. Thus, sub-blockrepeater 2002 generates block 2014, which comprises 2 blocks 2012.Sub-block scrambler 2004 is then configured to receive block 2014 and toscramble it, thus generating block 2016. One method of scrambling can beto invert half of block 2014 as illustrated in block 2016. But otherscrambling methods can also be implemented depending on the embodiment.

Sub-block terminator 2006 takes block 2016 generated by sub-blockscrambler 2004 and adds a termination block 2034 to the front of block2016 to form block 2018. Termination block 2034 ensures that each blockcan be processed independently in the receiver. Without terminationblock 2034, some blocks may be delayed due to multipath, for example,and they would therefore overlap part of the next block of data. But byincluding termination block 2034, the delayed block can be preventedfrom overlapping any of the actual data in the next block.

Termination block 2034 can be a cyclic prefix termination 2036. A cyclicprefix termination 2036 simply repeats the last few symbols of block2018. Thus, for example, if cyclic prefix termination 2036 is threesymbols long, then it would simply repeat the last three symbols ofblock 2018. Alternatively, termination block 2034 can comprise asequence of symbols that are known to both the transmitter and receiver.The selection of what type of block termination 2034 to use can impactwhat type of equalizer is used in the receiver. Therefore, receivercomplexity and choice of equalizers must be considered when determiningwhat type of termination block 2034 to use in TDM/FDM block 2000.

After sub-block terminator 2006, TDM/FDM block 2000 can include asub-block repeater 2008 configured to perform a second block repetitionstep in which block 2018 is repeated to form block 2020. In certainembodiments, sub-block repeater can be configured to perform a secondblock scrambling step as well. After sub-block repeater 2008, ifincluded, TDM/FDM block 2000 comprises a SYNC inserter 210 configured toperiodically insert an appropriate synchronization code 2032 after apredetermined number of blocks 2020 and/or to insert known symbols intoeach block. The purpose of synchronization code 2032 is discussed insection 3.

FIG. 21, on the other hand, illustrates an example embodiment of aTDM/FDM block 2100 configured for FDM, which comprises sub-blockrepeater 2102, sub-block scrambler 2104, block coder 2106, sub-blocktransformer 2108, sub-block terminator 2110, and SYNC inserter 2112.Sub-block repeater 2102 repeats block 2114 and generates block 2116.Sub-block scrambler then scrambles block 2116, generating block 2118.Sub-block coder 2106 takes block 2118 and codes it, generating block2120. Coding block correlates the data symbols together and generatessymbols b. This requires joint demodulation in the receiver, which ismore robust but also more complex. Sub-block transformer 2108 thenperforms a transformation on block 2120, generating block 2122.Preferably, the transformation is an IFFT of block 2120, which allowsfor more efficient equalizers to be used in the receiver. Next,sub-block terminator 2110 terminates block 2122, generating block 2124and SYNC inserter 2112 periodically inserts a synchronization code 2126after a certain number of blocks 2124 and/or insert known symbols intoeach block. Preferably, sub-block terminator 2110 only uses cyclicprefix termination as described above. Again this allows for moreefficient receiver designs.

TDM/FDM block 2100 is provided to illustrate the logical components thatcan be included in a TDM/FDM block configured to perform FDM on a datastream. Depending on the actual implementation, some of the logicalcomponents may or may not be included. Moreover, TDM/FDM block 2000 and2100 are preferably programmable so that the appropriate logicalcomponents can be included as required by a particular implementation.This allows a device that incorporates one of blocks 2000 or 2100 tomove between different systems with different requirements. Further, itis preferable that TDM/FDM block 1608 in FIG. 16 be programmable so thatit can be programmed to perform TDM, such as described in conjunctionwith block 2000, or FDM, such as described in conjunction with block2100, as required by a particular communication system.

After TDM/FDM blocks 1608, in FIG. 16, the parallel data streams arepreferably passed to interpolators 1610.

After interpolators 1610, the parallel data streams are passed tofilters 1612, which apply the pulse shapping described in conjunctionwith the roll-off factor of equation (2) in section 1. Then the paralleldata streams are sent to frequency shifter 1614, which is configured toshift each parallel data stream by the frequency offset associated withthe sub-channel to which the particular parallel data stream isassociated.

FIG. 22 illustrates an example embodiment of a frequency shifter 2200 inaccordance with the systems and methods described herein. As can beseen, frequency shifter 2200 comprises multipliers 2202 configured tomultiply each parallel data stream by the appropriate exponential toachieve the required frequency shift. Each exponential is of the form:exp(j2πf_(c)nT/rA), where c is the corresponding sub-channel, e.g., c=0to N−1, and n is time. Preferably, frequency shifter 1614 in FIG. 16 isprogrammable so that various channel/sub-channel configurations can beaccommodated for various different systems. Alternatively, an IFFT blockcan replace shifter 1614 and filtering can be done after the IFFT block.This type of implementation can be more efficient depending on theimplementation.

After the parallel data streams are shifted, they are summed, e.g., insummer 1512 of FIG. 15. The summed data stream is then transmitted usingthe entire bandwidth B of the communication channel being used. But thetransmitted data stream also comprises each of the parallel data streamsshifted in frequency such that they occupy the appropriate sub-channel.Thus, each sub-channel may be assigned to one user, or each sub-channelmay carry a data stream intended for different users. The assignment ofsub-channels is described in section 3 b. Regardless of how thesub-channels are assigned, however, each user will receive the entirebandwidth, comprising all the sub-channels, but will only decode thosesub-channels assigned to the user.

6. Sample Receiver Embodiments

FIG. 23 illustrates an example embodiment of a receiver 2300 that can beconfigured in accordance with the present invention. Receiver 2300comprises an antenna 2302 configured to receive a message transmitted bya transmitter, such as transmitter 1500. Thus, antenna 2302 isconfigured to receive a wide band message comprising the entirebandwidth B of a wide band channel that is divided into sub-channels ofbandwidth B. As described above, the wide band message comprises aplurality of messages each encoded onto each of a correspondingsub-channel. All of the sub-channels may or may not be assigned to adevice that includes receiver 2300; therefore, receiver 2300 may or maynot be required to decode all of the sub-channels.

After the message is received by antenna 2300, it is sent to radioreceiver 2304, which is configured to remove the carrier associated withthe wide band communication channel and extract a baseband signalcomprising the data stream transmitted by the transmitter. Example radioreceiver embodiments are described in more detail below.

The baseband signal is then sent to correlator 2306 and demodulator2308. Correlator 2306 is configured to correlated with a synchronizationcode inserted in the data stream as described in section 3. It is alsopreferably configured to perform SIR and multipath estimations asdescribed in section 3(b). Demodulator 2308 is configured to extract theparallel data streams from each sub-channel assigned to the devicecomprising receiver 2300 and to generate a single data stream therefrom.

FIG. 24 illustrates an example embodiment of a demodulator 2400 inaccordance with the systems and methods described herein. Demodulator2402 comprises a frequency shifter 2402, which is configured to apply afrequency offset to the baseband data stream so that parallel datastreams comprising the baseband data stream can be independentlyprocessed in receiver 2400. Thus, the output of frequency shifter 2402is a plurality of parallel data streams, which are then preferablyfiltered by filters 2404. Filters 2404 apply a filter to each paralleldata stream that corresponds to the pulse shape applied in thetransmitter, e.g., transmitter 1500. Alternatively, an IFFT block canreplace shifter 1614 and filtering can be done after the IFFT block.This type of implementation can be more efficient depending on theimplementation.

Next, receiver 2400 preferably includes decimators 2406 configured todecimate the data rate of the parallel bit streams. Sampling at higherrates helps to ensure accurate recreation of the data. But the higherthe data rate, the larger and more complex equalizer 2408 becomes. Thus,the sampling rate, and therefore the number of samples, can be reducedby decimators 2406 to an adequate level that allows for a smaller andless costly equalizer 2408.

Equalizer 2408 is configured to reduce the effects of multipath inreceiver 2300. Its operation will be discussed more fully below. Afterequalizer 2408, the parallel data streams are sent to de-scrambler,decoder, and de-interleaver 2410, which perform the opposite operationsof scrambler, encoder, and interleaver 1506 so as to reproduce theoriginal data generated in the transmitter. The parallel data streamsare then sent to parallel to serial converter 2412, which generates asingle serial data stream from the parallel data streams.

Equalizer 2408 uses the multipath estimates provided by correlator 2306to equalize the effects of multipath in receiver 2300. In oneembodiment, equalizer 2408 comprises Single-In Single-Out (SISO)equalizers operating on each parallel data stream in demodulator 2400.In this case, each SISO equalizer comprising equalizer 2408 receives asingle input and generates a single equalized output. Alternatively,each equalizer can be a Multiple-In Multiple-Out (MIMO) or a Multiple-InSingle-Out (MISO) equalizer. Multiple inputs can be required forexample, when a frequency encoder or rate controller, such as frequencyencoder 1900, is included in the transmitter. Because frequency encoder1900 encodes information from more than one parallel data stream ontoeach sub-channel, each equalizers comprising equalizer 2408 need toequalize more than one sub-channel. Thus, for example, if a paralleldata stream in demodulator 2400 comprises d(1)+d(8), then equalizer 2408will need to equalize both d(1) and d(8) together. Equalizer 2408 canthen generate a single output corresponding to d(1) or d(8) (MISO) or itcan generate both d(1) and d(8) (MIMO).

Equalizer 2408 can also be a time domain equalizer (TDE) or a frequencydomain equalizer (FDE) depending on the embodiment. Generally, equalizer2408 is a TDE if the modulator in the transmitter performs TDM on theparallel data streams, and a FDE if the modulator performs FDM. Butequalizer 2408 can be an FDE even if TDM is used in the transmitter.Therefore, the preferred equalizer type should be taken intoconsideration when deciding what type of block termination to use in thetransmitter. Because of power requirements, it is often preferable touse FDM on the forward link and TDM on the reverse link in a wirelesscommunication system.

As with transmitter 1500, the various components comprising demodulator2400 are preferably programmable, so that a single device can operate ina plurality of different systems and still maintain superiorperformance, which is a primary advantage of the systems and methodsdescribed herein. Accordingly, the above discussion provides systems andmethods for implementing a channel access protocol that allows thetransmitter and receiver hardware to be reprogrammed slightly dependingon the communication system.

Thus, when a device moves from one system to another, it preferablyreconfigures the hardware, i.e. transmitter and receiver, as requiredand switches to a protocol stack corresponding to the new system. Animportant part of reconfiguring the receiver is reconfiguring, orprogramming, the equalizer because multipath is a main problem for eachtype of system. The multipath, however, varies depending on the type ofsystem, which previously has meant that a different equalizer isrequired for different types of communication systems. The channelaccess protocol described in the preceding sections, however, allows forequalizers to be used that need only be reconfigured slightly foroperation in various systems.

a. Sample Equalizer Embodiment

FIG. 25 illustrates an example embodiment of a receiver 2500illustrating one way to configure equalizers 2506 in accordance with thesystems and methods described herein. Before discussing theconfiguration of receiver 2500, it should be noted that one way toconfigure equalizers 2506 is to simply include one equalizer per channel(for the systems and methods described herein, a channel is theequivalent of a sub-channel as described above). A correlator, such ascorrelator 2306 (FIG. 23), can then provide equalizers 2506 with anestimate of the number, amplitude, and phase of any multipaths present,up to some maximum number. This is also known as the Channel ImpulseResponse (CIR). The maximum number of multipaths is determined based ondesign criteria for a particular implementation. The more multipathsincluded in the CIR the more path diversity the receiver has and themore robust communication in the system will be. Path diversity isdiscussed a little more fully below.

If there is one equalizer 2506 per channel, the CIR is preferablyprovided directly to equalizers 2506 from the correlator (not shown). Ifsuch a correlator configuration is used, then equalizers 2506 can be runat a slow rate, but the overall equalization process is relatively fast.For systems with a relatively small number of channels, such aconfiguration is therefore preferable. The problem, however, is thatthere is large variances in the number of channels used in differenttypes of communication systems. For example, an outdoor system can havehas many as 256 channels. This would require 256 equalizers 2506, whichwould make the receiver design too complex and costly. Thus, for systemswith a lot of channels, the configuration illustrated in FIG. 25 ispreferable. In receiver 2500, multiple channels share each equalizer2506. For example, each equalizer can be shared by 4 channels, e.g.,CH1-Ch4, Ch5-CH8, etc., as illustrated in FIG. 25. In which case,receiver 2500 preferably comprises a memory 2502 configured to storeinformation arriving on each channel.

Memory 2502 is preferably divided into sub-sections 2504, which are eachconfigured to store information for a particular subset of channels.Information for each channel in each subset is then alternately sent tothe appropriate equalizer 2506, which equalizes the information based onthe CIR provided for that channel. In this case, each equalizer must runmuch faster than it would if there was simply one equalizer per channel.For example, equalizers 2506 would need to run 4 or more times as fastin order to effectively equalize 4 channels as opposed to 1. Inaddition, extra memory 2502 is required to buffer the channelinformation. But overall, the complexity of receiver 2500 is reduced,because there are fewer equalizers. This should also lower the overallcost to implement receiver 2500.

Preferably, memory 2502 and the number of channels that are sent to aparticular equalizer is programmable. In this way, receiver 2500 can bereconfigured for the most optimum operation for a given system. Thus, ifreceiver 2500 were moved from an outdoor system to an indoor system withfewer channels, then receiver 2500 can preferably be reconfigured sothat there are fewer, even as few as 1, channel per equalizer. The rateat which equalizers 2506 are run is also preferably programmable suchthat equalizers 2506 can be run at the optimum rate for the number ofchannels being equalized.

In addition, if each equalizer 2506 is equalizing multiple channels,then the CIR for those multiple paths must alternately be provided toeach equalizer 2506. Preferably, therefore, a memory (not shown) is alsoincluded to buffer the CIR information for each channel. The appropriateCIR information is then sent to each equalizer from the CIR memory (notshown) when the corresponding channel information is being equalized.The CIR memory (not shown) is also preferably programmable to ensureoptimum operation regardless of what type of system receiver 2500 isoperating in.

Returning to the issue of path diversity, the number of paths used byequalizers 2506 must account for the delay spread d_(s) in the system.For example, if the system is an outdoor system operating in the 5 GHzrange, the communication channel can comprise a bandwidth of 125 MHz,e.g., the channel can extend from 5.725 GHz to 5.85 GHz. If the channelis divided into 512 sub-channels with a roll-off factor r of 0.125, theneach sub-channel will have a bandwidth of approximately 215 KHz, whichprovides approximately a 4.6 μs symbol duration. Since the worst casedelay spread d_(s) is 20 μs, the number of paths used by equalizers 2504can be set to a maximum of 5. Thus, there would be a first path P1 at 0μs, a second path P2 at 4.6 μs, a third path P3 at 9.2 μs, a fourth pathP4 at 13.8 μs, and fifth path P5 at 18.4 μs, which is close to the delayspread d_(s).

In another embodiment, a sixth path can be included so as to completelycover the delay spread d_(s); however, 20 μs is the worst case. In fact,a delay spread d_(s) of 3 μs is a more typical value. In most instances,therefore, the delay spread d_(s) will actually be shorter and an extrapath is not needed. Alternatively, fewer sub-channels can be used, thusproviding a larger symbol duration, instead of using an extra path. Butagain, this would typically not be needed.

As explained above, equalizers 2506 are preferably configurable so thatthey can be reconfigured for various communication systems. Thus, forexample, the number of paths used must be sufficient regardless of thetype of communication system. But this is also dependent on the numberof sub-channels used. If, for example, receiver 2500 went from operatingin the above described outdoor system to an indoor system, where thedelay spread d_(s) is on the order of 1 μs, then receiver 2500 canpreferably be reconfigured for 32 sub-channels and 5 paths. Assuming thesame overall bandwidth of 125 MHz, the bandwidth of each sub-channel isapproximately 4 MHz and the symbol duration is approximately 250 ns.

Therefore, there will be a first path P1 at 0 μs and subsequent paths P2to P5 at 250 ns, 500 ns, 750 ns, and 1 μs, respectively. Thus, the delayspread d_(s) should be covered for the indoor environment. Again, the 1μs ds is worst case so the 1 us ds provided in the above example willoften be more than is actually required. This is preferable, however,for indoor systems, because it can allow operation to extend outside ofthe inside environment, e.g., just outside the building in which theinside environment operates. For campus style environments, where a useris likely to be traveling between buildings, this can be advantageous.

7. Sample Embodiment of a Wireless Communication Device

FIG. 26 illustrates an example embodiment of a wireless communicationdevice in accordance with the systems and methods described herein.Device 2600 is, for example, a portable communication device configuredfor operation in a plurality of indoor and outdoor communicationsystems. Thus, device 2600 comprises an antenna 2602 for transmittingand receiving wireless communication signals over a wirelesscommunication channel 2618. Duplexor 2604, or switch, can be included sothat transmitter 2606 and receiver 2608 can both use antenna 2602, whilebeing isolated from each other. Duplexors, or switches used for thispurpose, are well known and will not be explained herein.

Transmitter 2606 is a configurable transmitter configured to implementthe channel access protocol described above. Thus, transmitter 2606 iscapable of transmitting and encoding a wideband communication signalcomprising a plurality of sub-channels. Moreover, transmitter 2606 isconfigured such that the various subcomponents that comprise transmitter2606 can be reconfigured, or programmed, as described in section 5.Similarly, receiver 2608 is configured to implement the channel accessprotocol described above and is, therefore, also configured such thatthe various sub-components comprising receiver 2608 can be reconfigured,or reprogrammed, as described in section 6.

Transmitter 2606 and receiver 2608 are interfaced with processor 2610,which can comprise various processing, controller, and/or Digital SignalProcessing (DSP) circuits. Processor 2610 controls the operation ofdevice 2600 including encoding signals to be transmitted by transmitter2606 and decoding signals received by receiver 2608. Device 2610 canalso include memory 2612, which can be configured to store operatinginstructions, e.g., firmware/software, used by processor 2610 to controlthe operation of device 2600.

Processor 2610 is also preferably configured to reprogram transmitter2606 and receiver 2608 via control interfaces 2614 and 2616,respectively, as required by the wireless communication system in whichdevice 2600 is operating. Thus, for example, device 2600 can beconfigured to periodically ascertain the availability is a preferredcommunication system. If the system is detected, then processor 2610 canbe configured to load the corresponding operating instruction frommemory 2612 and reconfigure transmitter 2606 and receiver 2608 foroperation in the preferred system.

For example, it may preferable for device 2600 to switch to an indoorwireless LAN if it is available. So device 2600 may be operating in awireless WAN where no wireless LAN is available, while periodicallysearching for the availability of an appropriate wireless LAN. Once thewireless LAN is detected, processor 2610 will load the operatinginstructions, e.g., the appropriate protocol stack, for the wireless LANenvironment and will reprogram transmitter 2606 and receiver 2608accordingly. In this manner, device 2600 can move from one type ofcommunication system to another, while maintaining superior performance.

It should be noted that a base station configured in accordance with thesystems and methods herein will operate in a similar manner as device2600; however, because the base station does not move from one type ofsystem to another, there is generally no need to configure processor2610 to reconfigure transmitter 2606 and receiver 2608 for operation inaccordance with the operating instruction for a different type ofsystem. But processor 2610 can still be configured to reconfigure, orreprogram the sub-components of transmitter 2606 and/or receiver 2608 asrequired by the operating conditions within the system as reported bycommunication devices in communication with the base station. Moreover,such a base station can be configured in accordance with the systems andmethods described herein to implement more than one mode of operation.In which case, controller 2610 can be configured to reprogramtransmitter 2606 and receiver 2608 to implement the appropriate mode ofoperation.

8. Bandwidth Recovery

As described above in relation to FIGS. 11-14, when a device, such asdevice 1118 is near the edge of a communication cell 1106, it mayexperience interference from base station 1112 of an adjacentcommunication cell 1104. In this case, device 1118 will report a low SIRto base station 1114, which will cause base station 1114 to reduce thenumber of sub-channels assigned to device 1118. As explained in relationto FIGS. 12 and 13, this reduction can comprise base station 1114assigning only even sub-channels to device 1118. Preferably, basestation 1112 is correspondingly assigning only odd sub-channels todevice 1116.

In this manner, base station 1112 and 1114 perform complementaryreductions in the channels assigned to devices 1116 and 1118 in order toprevent interference and improve performance of devices 1116 and 1118.The reduction in assigned channels reduces the overall bandwidthavailable to devices 1116 and 1118. But as described above, a systemimplementing such a complementary reduction of sub-channels will stillmaintain a higher bandwidth than conventional systems. Still, it ispreferable to recover the unused sub-channels, or unused bandwidth,created by the reduction of sub-channels in response to a low reportedSIR.

One method for recovering the unused bandwidth is illustrated in theflow chart of FIG. 27. First, in step 2702, base station 1114 receivesSIR reports for different groups of sub-channels from device 1118 asdescribed above. If the group SIR reports are good, then base station1114 can assign all sub-channels to device 1118 in step 2704. If,however, some of the group SIR reports received in step 2702 are poor,then base station 1114 can reduce the number of sub-channels assigned todevice 1118, e.g., by assigning only even sub-channels, in step 2706. Atthe same time, base station 1112 is preferably performing acomplementary reduction in the sub-channels assigned to device 1116,e.g., by assigning only odd sub-channels.

At this point, each base station has unused bandwidth with respect todevices 1116 and 1118. To recover this bandwidth, base station 1114 can,in step 2708, assign the unused odd sub-channels to device 1116 inadjacent cell 1104. It should be noted that even though cells 1102,1104, and 1106 are illustrated as geometrically shaped, non-overlappingcoverage areas, the actual coverage areas do not resemble these shapes.The shapes are essentially fictions used to plan and describe a wirelesscommunication system 1100. Therefore, base station 1114 can in factcommunicate with device 1116, even though it is in adjacent cell 1104.

Once base station 1114 has assigned the odd sub-channels to device 1116,in step 2708, base station 1112 and 1114 communicate with device 1116simultaneously over the odd sub-channels in step 2710. Preferably, basestation 1112 also assigns the unused even sub-channels to device 1118 inorder to recover the unused bandwidth in cell 1104 as well.

In essence, spatial diversity is achieved by having both base station1114 and 1112 communicate with device 1116 (and 1118) over the samesub-channels. Spatial diversity occurs when the same message istransmitted simultaneously over statistically independent communicationpaths to the same receiver. The independence of the two paths improvesthe overall immunity of the system to fading. This is because the twopaths will experience different fading effects. Therefore, if thereceiver cannot receive the signal over one path due to fading, then itwill probably still be able to receive the signal over the other path,because the fading that effected the first path will not effect thesecond. As a result, spatial diversity improves overall systemperformance by improving the Bit Error Rate (BER) in the receiver, whicheffectively increases the deliverable data rate to the receiver, i.e.,increase the bandwidth.

For effective spatial diversity, base stations 1112 and 1114 ideallytransmit the same information at the same time over the samesub-channels. As mentioned above, each base station in system 1100 isconfigured to transmit simultaneously, i.e., system 1100 is a TDM systemwith synchronized base stations. Base stations 1112 and 1114 alsoassigned the same sub-channels to device 1116 in step 2708. Therefore,all that is left is to ensure that base stations 1112 and 1114 send thesame information. Accordingly, the information communicated to device1116 by base stations 1112 and 1114 is preferably coordinated so thatthe same information is transmitted at the same time. The mechanism forenabling this coordination is discussed more fully below. Suchcoordination, however, also allows encoding that can provide furtherperformance enhancements within system 1100 and allow a greaterpercentage of the unused bandwidth to be recovered.

One example coordinated encoding scheme that can be implemented betweenbase stations 1112 and 1114 with respect to communications with device1116 is Space-Time-Coding (STC) diversity. STC is illustrated by system2800 in FIG. 28. In system 2800, transmitter 2802 transmits a messageover channel 2808 to receiver 2806. Simultaneously, transmitter 2804transmits a message over channel 2810 to receiver 2806. Because channels2808 and 2810 are independent, system 2800 will have spatial diversitywith respect to communications from transmitters 2802 and 2804 toreceiver 2806. In addition, however, the data transmitted by eachtransmitter 2802 and 2804 can be encoded to also provide time diversity.The following equations illustrate the process of encoding and decodingdata in a STC system, such as system 2800.

First, channel 2808 can be denoted h_(n) and channel 2810 can be denotedg_(n), where:h _(n) =a _(h) e ^(jθh); and  (1)g _(n) =a _(g) e ^(jθg).  (2)

Second, we can look at two blocks of data 2812 a and 2812 b to betransmitted by transmitter 2802 as illustrated in FIG. 28. Block 2812 acomprises N-symbols denoted as a₀, a₁, a₂, . . . , a_(N−1), or a(0:N−1).Block 2812 b transmits N-symbols of data denoted b(0: N−1). Transmitter2804 simultaneously transmits two block of data 2814 a and 2814 b. Block2814 a is the negative inverse conjugate of block 2812 b and cantherefore be described as −b*(N−1:0). Block 2814's the inverse conjugateof block 2812 a and can therefore be described as a*(N−1:0). It shouldbe noted that each block of data in the forgoing description willpreferably comprise a cyclical prefix as described above.

When blocks 2812 a, 2812 b, 2814 a, and 2814 b are received in receiver2806, they are combined and decoded in the following manner: First, theblocks will be combined in the receiver to form the following blocks,after discarding the cyclical prefix:Block1=a(0:N−1){circle around (x)}h _(n) −b*(N−1:0){circle around (x)}g_(n); and  (3)Block2=b(0:−N−1){circle around (x)}h _(n) +a*(N−1:0){circle around (x)}g_(n).  (4)

Where the symbol {circle around (x)} represents a cyclic convolution.

Second, by taking an IFFT of the blocks, the blocks can be described as:Block1=A _(n) *H _(n) −B _(n) **G _(n); and  (5)Block2=B _(n) *H _(n) −A _(n) **G _(n).  (6)

Where n=0 to N−1.

In equations (5) and (6) H_(n) and G_(n) will be known, or can beestimated. But to solve the two equations and determine A_(n) and B_(n),it is preferable to turn equations (5) and (6) into two equations withtwo unknowns. This can be achieved using estimated signals X_(n) andY_(n) as follows:X _(n) =A _(n) *H _(n) −B _(n) **G _(n); and  (7)Y _(n) =B _(n) *H _(n) +A _(n) **G _(n).  (8)

To generate two equations and two unknowns, the conjugate of Y_(n) canbe used to generate the following two equations:X _(n) =A _(n) *H _(n) −B _(n) **G _(n); and  (9)Y _(n) *=B _(n) **H _(n) *+A _(n) *G _(n)*.  (10)

Thus, the two unknowns are A_(n) and B_(n)* and equations (9) and (10)define a matrix relationship in terms of these two unknowns as follows:

$\begin{matrix}{\begin{bmatrix}X_{n} \\Y_{n}^{*}\end{bmatrix} = {\begin{bmatrix}{H_{n} - G_{n}} \\{G_{n}^{*}H_{n}^{*}}\end{bmatrix}*\begin{bmatrix}A_{n} \\B_{n}^{*}\end{bmatrix}}} & (11)\end{matrix}$Which can be rewritten as:

$\begin{matrix}{\begin{bmatrix}A_{n} \\B_{n}^{*}\end{bmatrix} = {\frac{1}{H_{n}❘^{2}{+ {❘{G_{n}❘^{2}}}}}*\begin{bmatrix}{H_{n}^{*}G_{n}} \\{{- G_{n}^{*}}H_{n}}\end{bmatrix}*\begin{bmatrix}X_{n} \\Y_{n}^{*}\end{bmatrix}}} & (12)\end{matrix}$

Signals A_(n) and B_(n) can be determined using equation (12). It shouldbe noted, that the process just described is not the only way toimplement STC. Other methods can also be implemented in accordance withthe systems and methods described herein. Importantly, however, byadding time diversity, such as described in the preceding equations, tothe space diversity already achieved by using base stations 1112 and1114 to communicate with device 1116 simultaneously, the BER can bereduced even further to recover even more bandwidth.

An example transmitter 2900 configured to communicate using STC inaccordance with the systems and methods described herein is illustratedin FIG. 29. Transmitter 2900 includes a block storage device 2902, aserial-to-parallel converter 2904, encoder 2906, and antenna 2908. Blockstorage device 2902 is included in transmitter 2900 because a 1 blockdelay is necessary to implement the coding illustrated in FIG. 28. Thisis because transmitter 2804 first transmits −b_(n)*(n=N−1 to 0). Butb_(n) is the second block, so if transmitter 2900 is going to transmit−b_(n)* first, it must store two blocks, e.g., a_(n) and b_(n), and thengenerate block 2814 a and 2814 b (see FIG. 28).

Serial-to-parallel converter 2904 generates parallel bit streams fromthe bits of blocks a_(n) and b_(n). Encoder 2906 then encodes the bitstreams as required, e.g., encoder 2906 can generate −b_(n)* and a_(n)*(see blocks 2814 a and 2814 b in FIG. 28). The encoded blocks are thencombined into a single transmit signal as described above andtransmitted via antenna 2908.

Transmitter 2900 preferably uses TDM to transmit messages to receiver2806. An alternative transmitter 3000 embodiment that uses FDM isillustrated in FIG. 30. Transmitter 3000 also includes block storagedevice 3002, a serial-to-parallel converter 3004, encoder 3006, andantenna 3008, which are configured to perform in the same manner as thecorresponding components in transmitter 2900. But in addition,transmitter 3000 includes IFFTs 3010 to take the IFFT of the blocksgenerated by encoder 2906. Thus, transmitter 3000 transmits −B_(n)* andA_(n)* as opposed to −b_(n)* and a_(n)*, which provides space,frequency, and time diversity.

FIG. 31 illustrates an alternative system 3100 that also uses FDM butthat eliminates the 1 block delay associated with transmitters 2900 and3000. In system 3100, transmitter 3102 transmits over channel 3112 toreceiver 3116. Transmitter 3106 transmits over channel 3114 to receiver3116. As with transmitters 2802 and 2804, transmitters 3102 and 3106implement an encoding scheme designed to recover bandwidth in system3100. In system 3100, however, the coordinated encoding occurs at thesymbol level instead of the block level.

Thus, for example, transmitter 3102 can transmit block 3104 comprisingsymbols a₀, a₁, a₂, and a₃. In which case, transmitter 3106 willtransmit a block 3108 comprising symbols −a₁*, a₀* −a₃*, and a₂*. As canbe seen, this is the same encoding scheme used by transmitters 2802 and2804, but implemented at the symbol level instead of the block level. Assuch, there is no need to delay one block before transmitting. An IFFTof each block 3104 and 3108 can then be taken and transmitted using FDM.An IFFT 3110 of block 3104 is shown in FIG. 31 for purposes ofillustration.

Channels 3112 and 3114 can be described by H_(n) and G_(n),respectively. Thus, in receiver 3116 the following symbols will beformed:(A₀*H₀)−(A₁**G₀)(A₁*H₁)+(A₀**G₁)(A₂*H₂)−(A₃**G₂)(A₃*H₃)+(A₂**G₃).

In time, each symbol a_(n) (n=0 to 3) occupies a slightly different timelocation. In frequency, each symbol A_(n) (n=0 to 3) occupies a slightlydifferent frequency. Thus, each symbol A_(n) is transmitted over aslightly different channel, i.e., H_(n) (n=0 to 3) or G_(n) (n=0 to 3),which results in the combinations above.

As can be seen, the symbol combinations formed in the receiver are ofthe same form as equations (5) and (6) and, therefore, can be solved inthe same manner, but without the one block delay.

In order to implement STC or Space Frequency Coding (SFC) diversity asdescribed above, bases stations 1112 and 1114 must be able to coordinateencoding of the symbols that are simultaneously sent to a particulardevice, such as device 1116 or 1118. Fortunately, base stations 1112 and1114 are preferably interfaced with a common network interface server.For example, in a LAN, base stations 1112 and 1114 (which would actuallybe service access points in the case of a LAN) are interfaced with acommon network interface server that connects the LAN to a largernetwork such as a Public Switched Telephone Network (PSTN). Similarly,in a wireless WAN, base stations 1112 and 1114 are typically interfacedwith a common base station control center or mobile switching center.Thus, coordination of the encoding can be enabled via the commonconnection with the network interface server. Bases station 1112 and1114 can then be configured to share information through this commonconnection related to communications with devices at the edge of cells1104 and 1106. The sharing of information, in turn, allows time orfrequency diversity coding as described above.

It should be noted that other forms of diversity, such as polarizationdiversity or delay diversity, can also be combined with the spatialdiversity in a communication system designed in accordance with thesystems and methods described herein. The goal being to combinealternative forms of diversity with the spatial diversity in order torecover larger amounts of bandwidth. It should also be noted, that thesystems and methods described can be applied regardless of he number ofbase stations, devices, and communication cells involved.

Briefly, delay diversity can preferably be achieved in accordance withthe systems and methods described herein by cyclical shifting thetransmitted blocks. For example, one transmitter can transmit a blockcomprising A₀, A₁, A₂, and A₃ in that order, while the other transmittertransmits the symbols in the following order A₃, A₀, A₁, and A₂.Therefore, it can be seen that the second transmitter transmits acyclically shifted version of the block transmitted by the firsttransmitter. Further, the shifted block can be cyclically shifted bymore then one symbol of required by a particular implementation.

9. Transmitter Radio Module

FIG. 32 is a diagram illustrating a conventional radio transmittermodule 3200. In certain embodiments, such a conventional radiotransmitter module can, for example, be used to implement radiotransmitter module 1514. As can be seen, baseband circuitry 3202 can beconfigured to provide a digital transmit signal to radio transmittermodule 3200 for transmission. Baseband circuitry 3202 can, for example,comprise the components described above in relation to FIGS. 15-23.

The digital transmit data provided by baseband circuitry 3202 can beseparated into a plurality of data streams, such as the Inphase (I) andQuadrature phase (Q) data streams illustrated in FIG. 32. The I and Qdata streams can then be encoded onto two orthogonal waveforms. Thedivision of baseband data stream 1518 to I and Q channels can take on avariety of schemes. For example, if data stream 1518 is complex, thenthe I-channel can represent the real part of data stream 1518 and theQ-channel can represent the imaginary part of data stream 1518.Separation of digital transmit data into I and Q data streams is wellknown and will not be discussed in further detail here.

In many conventional radio transmit modules, the digital I and Q datastreams are converted to analog signals by digital to analog (D/A)converters 3204 and 3254 respectively. The resultant analog signals caneach be filtered with low-pass filters 3206 and 3256, respectively. Thefiltered signals can then be modulated by modulators 3208 and 3258 witha local oscillator (LO) signal centered at the carrier frequency ω₀. Asynthesizer 3210 coupled to a local oscillator 3210 can be used togenerated the LO signals used by both mixers 3208 and 3258. Themodulated signals supplied by the mixers can be combined by combiner3212. The combined signal can be filtered through band pass filter 3214.The filtered signal is then amplified by power amplifier 3216 andbroadcast with antenna 3218.

The implementation illustrated in FIG. 32 is commonly referred to as adirect conversion transmitter, because the baseband signals areconverted directly to a signal residing at the RF carrier frequency ω₀.In other embodiments, a staged approach to frequency conversion in whichthe baseband signal is first stepped up to one pr more intermediatefrequencies before being converted to an RF signal can be implemented.Such multi-staged transmitters often comprise additional mixers,synthesizers, local oscillators, etc.

The synthesizers, local oscillators, and D/A converters required byconventional radio transmit modules can be large and expensive and canhave relative large power requirements, especially when they are run athigher data rates. As communication data rates for new systems increase,the power consumption required by such components can be prohibitive.

FIG. 33 illustrates an example embodiment of a radio transmit module3300 configured in accordance with the systems and methods describedherein. Unlike radio transmit module 3200, radio transmit module 3300does not include D/As, synthesizers, local oscillators or modulators. Asa result, radio transmit module 3300 can avoid the expense, sizeconstraints, and power constraints that are inherent in conventionalradio transmit module designs.

In order to achieve such benefits, baseband circuitry 3202 can, forexample, be configured to use 3 signal levels: “1”, “0”, and “−1” forboth the I and Q data. By using two signals for each data stream,however, radio transmit module 3300 only sees a series of 1s and 0s. Forexample, as illustrated in FIG. 34, the I+ data stream can be coded suchthat it goes high when a “1” is being transmitted and stays low wheneither a “0” or a “−1” is being transmitted. The I− data stream can becoded such that it goes high when a “−1” is being transmitted and stayslow when either a “0” or a “1” is being transmitted. The Q+ and Q− datastreams can be coded in the same manner. The data streams can then bepassed through pulsers 3322-3328, which can be configured to convert thedata bits in each data stream into narrower pulses. The narrow pulsescan then be combined in combiners 3320 and 3322 such that:I=(I+)−(I−); and  (1)Q=(Q+)−(Q−).  (2)

The combined signals can then be band pass filtered using band passfilters 3334 and 3336, which in addition to confining the signal to anarrow bandwidth for transmission, can also shape the pulse sequences ineffect modulating the signals. The resultant shaped signals are thenready for transmission in the appropriate frequency range. The shapedsignals can then be combined in adder 3338, amplified by amplifier 3340,and transmitted via antenna 3342. Band-pass filters 3334 and 3336 canalso be configured to control phase. In order to maintain orthogonalwaveforms, band-pass filter 3334 can be configured to only allow sin ω₀tcomponents to pass and band-pass filter 3336 can be configured to allowcos ω₀t components to pass which are orthogonal waveforms.

It should be noted that, in certain embodiments, amplifier 3340 can bereplaced by an appropriate driver.

Thus, by constraining the digital data streams to values of “1”, “0”,and “−1” radio transmitter module 3300 can be implemented withoutcostly, power consuming D/As, synthesizers, modulators/mixers, etc.Accordingly, radio transmit modules configured in accordance with thesystems and methods described herein can provide cost, size, and powerbenefits that cannot be achieved by conventional transmitter designs.

FIG. 35 is a diagram illustrating an example embodiment of a pulserconfigured in accordance with the systems and methods described herein.In the example of FIG. 35, the pulser comprises an AND gate 3506 and adelay module 3504. A data stream 3502 is coupled to one input of ANDgate 3506 as well as to the delay module 3504, which produces a delayedversion 3508 of data stream 3502. Delayed data stream 3508 is thencoupled to the other input of AND gate 3506. As can be seen on the rightside of FIG. 35, when data stream 3502 is ANDed with delayed data stream3508, a data stream 3510 comprising narrower pulses can be created. Theamount of delay applied by delay module 3504 must, of course, becontrolled so as to produce satisfactory pulse widths in data stream3510.

As illustrated in FIG. 35, an inverted output can be used to generateoutputs for the I− and Q− data streams so that they can be combined withthe I+ and Q+ data streams in accordance with equations (1) and (2). Insuch embodiments, combiners 3320 and 3322 can be passive combiners suchas the one illustrated in FIG. 36. The combiner illustrated in FIG. 36simply comprises three resistive components R1, R2, and R3. Theresistive value of components R1, R2, and R3 can be selected on animplementation by implementation bases. For example, in certainembodiments the resistive values can be selected such that R1=R2=R3. Forexample, a value of 17 ohms can be used for each of R1, R2, and R3.

FIG. 37 illustrates an alternative embodiment of a pulser that can beused to implement pulsers 3322-3328. In this embodiment, a capacitor C1and resistor R4 in what can be referred to as an edge detectorconfiguration are used. When a data stream 3708 is passed through thecapacitor-resistor combination, the rising and falling edges of thevarious bits will create narrower positive and negative pulses asillustrated by waveform 3710. In order to eliminate the negative pulses,a diode D1 can be included so that waveform 3710 is converted intowaveform 3712, which comprises a narrow pulse for each “1” in datastream 3708.

The decay rate of the pulse comprising waveform 3712 is determined bythe capacitance of capacitor C1 and the resistance of resistor R1. Toproduce narrow pulses, the circuit decay rate 1/RC should be much lessthan the data rate. The pulser of FIG. 37 can also be referred to as adifferentiator detector. For a differentiator detector to work, theinput data stream must adhere to a “return to zero” convention, that isregardless of the value of a datum the signal always returns to zerobefore the transmission of the next datum.

When pulsers such as the one illustrated in FIG. 37 are used, an activecombiner can be used to combine I+ and Q+ with I− and Q−, respectively,in accordance with equations (1) and (2). An active combiner cancomprise an Operational Amplifier (OP-AMP) 3802 as illustrated in FIG.38. I+ or Q+ can be coupled to the positive input of OP-AMP 3802, whileI− or Q− can be coupled to the negative input.

10. Radio Receiver

FIG. 39 is diagram illustrating an exemplary radio receiver 3900 thatcan be used, for example, to implement radio receiver 2304. As can beseen, a radio signal is first received by antenna 3902 and filtered byband-pass filter 3904. The filtered signal can then be amplified byamplifier 3906, which can comprises a low noise amplifier (LNA) and insome embodiments include additional amplifiers. The amplified signal canthen be split into I and Q components and down converted by mixers 3910and 3960 which have an LO signal supplied to them by the combination ofsynthesizer 3908 and local oscillator 3970.

Once the signals are downconverted, low pass filters 3912 and 3962 canbe configured to eliminate high frequency artifacts from thedownconverted signals. The resultant filter signals can then, forexample, be converted to digital signals by analog-to-digital (A/D)converters 3914 and 3964. The digital I and Q signals can then besupplied to baseband circuit 3920 for further processing. In someimplementations, additional mixers, and frequency filters can beemployed to perform the downconversion in several steps, for exampledownconverting from RF to IF and then downconverting from IF to basebandfrequency.

As with the radio transmitter module of FIG. 32, it is well known thatthe synthesizer, local oscillator, mixers, and A/D converters can belarge, expensive, and consume a relatively large amount of power. If atransmitter such as the one illustrated in FIG. 33 is used on thetransmit side, however, then a corresponding radio receiver can be usedthat eliminates the need for such components in the receiver. FIG. 40 isa diagram of an example radio receiver 4000 that can be configured towork, for example, with the radio transmit module of FIG. 33. In radioreceiver 4000, a signal is received by antenna 4002, filtered byband-pass filter 4004, and amplified by amplifier 4006, in a mannersimilar to that of the receiver described in FIG. 39.

The signal can then be processed in two concurrent processes. In thefirst process, the envelope, for example, of the signal can be detectedusing detector 4010. Depending on the embodiment, for example, detector4010 can be an envelop detector or a power detector. Envelope detectorsare well known and can, for example, be implemented as a simple diode ora triode with the proper biasing. The output of envelop detector 4010can then be filtered by filter 4012 and converted to a digital signal.Filter 4012 can be implemented as a low pass filter with DC removal,e.g., a single pole notch filter. The conversion process can beachieved, for example, using A/D converter 4014, which then forwards thedigital signal to base band circuitry 4018.

Sign detector 4020 can then be used to detect the sign of the bit beingdecoded. Thus, when the output of A/D converter 4014 is combined withthe output of sign detector 4020, the original “1”, “0”, “−1” values canbe recovered by baseband circuitry 4018. Sign detection can, dependingon the embodiment, be implemented using a limiter and circuitryconfigured to detect a double positive, or double negative, in theresulting bit stream.

FIG. 41 is a diagram illustrating an alternative embodiment of a radioreceiver 4100 that can be used in conjunction, for example, with thetransmitter of FIG. 33. In radio receiver 4100, an sigma-delta A/Dconverter is formed by combiner 4102, band pass filter 4104, precision,clocked comparator 4110, and D/A converter 4106. Thus, the incomingsignal is filtered by filter 4104 and then compared to a groundreference by comparator 4104. The output of comparator 4110 is then fedto D/A converter 4106, the output of which is then subtracted form theincoming signal by combiner 4102. The output if comparator 4110 is alsosent to filter and decimation circuitry 4108, the output of which issent to baseband circuitry 4112.

The incoming signal is over sampled. The over sampling factor and orderof band pass filter 4104 result in a certain effective number of bits atthe output of filtering and decimation circuitry 4108. The table in FIG.42 illustrates the effective number of bits for each sampling frequency,i.e., the rate at which comparator 4110 is clocked, the filter order forband pass filter 4104, and the over sampling factor, for a particularimplementation.

In an alternative embodiment, over sampling can be achieved using aplurality of comparators, each clocked on a different phase of a clocksignal. The output of the comparators can then be combined and passed toD/A converter 4106 and filtering a decimation circuitry 4108.

While embodiments and implementations of the invention have been shownand described, it should be apparent that many more embodiments andimplementations are within the scope of the invention. Accordingly, theinvention is not to be restricted, except in light of the claims andtheir equivalents.

1. A radio transmitter module configured to generate a modulated signalhaving a three-level signal constellation from a data stream comprisinga plurality of symbols, each symbol having one of three symbol values,the transmitter, comprising: a first pulser configured for generating ahigh signal for each of the plurality of symbols having a first symbolvalue and a low signal for each of the plurality of symbols having asecond symbol value or a third symbol value; a second pulser configuredfor generating a high signal for each of the plurality of symbols havingthe third symbol value and a low signal for each of the plurality ofsymbols having the second symbol value or the first symbol value, thesecond pulser further comprising an inverted output; and a combinercoupled to the first pulser and the second pulser for combining pulsedsignal outputs to generate the modulated signal.
 2. The radiotransmitter module of claim 1, wherein the three symbol values comprise“1”, “0”, and “−1”.
 3. The radio transmitter module of claim 1, furthercomprising a pulse-shaping filter for filtering the pulsed signaloutputs.
 4. The radio transmitter module of claim 1, wherein thecombiner is configured to combine the pulsed signal outputs bysubtracting signals generated by the second pulser from signalsgenerated by the first pulser.
 5. The radio transmitter module of claim1, wherein the modulated signal produced by the first pulser, the secondpulser, and the combiner comprises an in-phase signal, the radiotransmitter further comprising: a third pulser configured for generatinga high signal for each of a plurality of input symbols having the firstsymbol value and a low signal for each of the plurality of input symbolshaving the second symbol value or the third symbol value; a fourthpulser configured for generating a high signal for each of the pluralityof input symbols having the third symbol value and a low signal for eachof the plurality of input symbols having the second symbol value or thefirst symbol value, the second pulser further comprising an invertedoutput; combiner coupled to the third pulser and the fourth pulser, thecombiner configured to combine pulsed outputs generated by the thirdpulser and the fourth pulser for producing a quadrature-phase signal. 6.The radio transmitter module of claim 5, wherein the three symbol valuescomprise “1”, “0”, and “−1”.
 7. The radio transmitter module of claim 5,further comprising a pulse-shaping filter configured for filtering atleast one of a set of signals, the set comprising the pulsed outputs,the in-phase signal, and the quadrature-phase signal.
 8. The radiotransmitter module of claim 1, wherein the combiner is configured tocombine the pulsed signal outputs by subtracting signals generated bythe fourth pulser from signals generated by the fourth pulser.
 9. Theradio transmitter module of claim 5, further comprising an adderconfigured to combine the in-phase signal with the quadrature-phasesignal.
 10. The radio transmitter module of claim 5, further configuredfor encoding the in-phase signal onto a first waveform and encoding thequadrature-phase signal onto a second waveform, the first waveform beingorthogonal to the second waveform.
 11. The radio transmitter module ofclaim 1, wherein each of the first pulser and the second pulsercomprises an AND gate and a delay module.
 12. The radio transmittermodule of claim 11, wherein the output of the AND gate associated withthe second pulser is inverted.
 13. The radio transmitter module of claim5, wherein each of the third pulser and the fourth pulser comprises anAND gate and a delay module.
 14. The radio transmitter module of claim13, wherein the output of the AND gate associated with the fourth pulseris inverted.
 15. The radio transmitter module of claim 1, wherein eachof the first pulser and the second pulser comprises an edge detector.16. The radio transmitter module of claim 5, wherein each of the thirdpulser and the fourth pulser comprises an edge detector.
 17. The radiotransmitter module of claim 5, wherein the combiner comprises at leastone passive combiner.
 18. The radio transmitter module of claim 5,wherein the combiner comprises at least one active combiner.
 19. Atransmitter configured for receiving a data stream from a basebandcircuit and generating a modulated signal having a three-level signalconstellation, the transmitter comprising: a first pulser configured forgenerating a high signal for each of the plurality of symbols having afirst symbol value and a low signal for each of the plurality of symbolshaving a second symbol value or a third symbol value; a second pulserconfigured for generating a high signal for each of the plurality ofsymbols having the third symbol value and a low signal for each of theplurality of symbols having the second symbol value or the first symbolvalue; and a first combiner coupled to the first pulser and the secondpulser for subtracting a pulsed signal output of the second pulser froma pulsed signal output of the first pulser to generate the modulatedsignal.
 20. The transmitter of claim 19, wherein the three symbol valuescomprise “1”, “0”, and “−1”.
 21. The transmitter of claim 19, furthercomprising a pulse-shaping filter for filtering the pulsed signaloutputs.
 22. The transmitter of claim 19, wherein the second pulserfurther comprises an inverted output for generating an inverted pulsedoutput, the first combiner configured to sum the pulsed signal outputgenerated by the first pulser with the inverted pulsed output generatedby the second pulser.
 23. The transmitter of claim 19 wherein themodulated signal produced by the first pulser, the second pulser, andthe first combiner comprises an in-phase signal, the radio transmitterfurther comprising: a third pulser configured for generating a highsignal for each of a plurality of input symbols having the first symbolvalue and a low signal for each of the plurality of input symbols havingthe second symbol value or the third symbol value; a fourth pulserconfigured for generating a high signal for each of the plurality ofinput symbols having the third symbol value and a low signal for each ofthe plurality of input symbols having the second symbol value or thefirst symbol value; and a second combiner coupled to the third pulserand the fourth pulser, the second combiner configured to subtract apulsed output generated by the fourth pulser from a pulsed outputgenerated by the third pulser for producing a quadrature-phase signal.24. The transmitter of claim 23, wherein the three symbol valuescomprise “1”, “0”, and “−1”.
 25. The transmitter module of claim 23,further comprising a pulse-shaping filter configured for filtering atleast one of a set of signals, the set comprising the pulsed outputs,the in-phase signal, and the quadrature-phase signal.
 26. Thetransmitter of claim 23, wherein the fourth pulser further comprises aninverted output for generating an inverted pulsed output, the secondcombiner configured to sum the pulsed output generated by the thirdpulser with the inverted pulsed output generated by the fourth pulser.27. The transmitter of claim 23, further comprises an adder configuredto combine the in-phase signal with the quadrature-phase signal.
 28. Thetransmitter of claim 23, further configured for encoding the in-phasesignal onto a first waveform and encoding the quadrature-phase signalonto a second waveform, the first waveform being orthogonal to thesecond waveform.
 29. The transmitter of claim 19, wherein each of thefirst pulser and the second pulser comprises an AND gate and a delaymodule.
 30. The transmitter of claim 29, wherein the output of the ANDgate associated with the second pulser is inverted.
 31. The transmitterof claim 23, wherein each of the third pulser and the fourth pulsercomprises an AND gate and a delay module.
 32. The transmitter of claim23, wherein the output of the AND gate associated with the fourth pulseris inverted.
 33. The transmitter of claim 19, wherein each of the firstpulser and the second pulser comprises an edge detector.
 34. Thetransmitter of claim 23, wherein each of the third pulser and the fourthpulser comprises an edge detector.
 35. The transmitter of claim 23,wherein the first combiner and the second combiner comprise passivecombiners.
 36. The transmitter of claim 23, wherein the third combinerand the fourth combiner comprise active combiners.
 37. A method oftransmitting data in a wireless communication network, comprising:generating a high signal for each of the plurality of symbols having afirst symbol value and a low signal for each of the plurality of symbolshaving a second symbol value or a third symbol value for producing afirst pulsed signal output; generating a high signal for each of theplurality of symbols having the third symbol value and a low signal foreach of the plurality of symbols having the second symbol value or thefirst symbol value for producing a second pulsed signal output; andcombining the first pulsed signal output with an inverted version of thesecond pulsed signal output to generate at least one combined waveformhaving a three-level signal constellation.
 38. The method of claim 37,further comprising pulse shaping the at least one combined waveform. 39.The method of claim 37, wherein generating the high signal for each ofthe plurality of symbols having the first symbol value and the lowsignal for each of the plurality of symbols having the second symbolvalue or the third symbol value for producing the first pulsed signaloutput; and generating the high signal for each of the plurality ofsymbols having the third symbol value and the low signal for each of theplurality of symbols having the second symbol value or the first symbolvalue for producing the second pulsed signal output comprises ANDingeach of a plurality of data streams with a delayed version of itself.40. The method of claim 37, wherein generating the high signal for eachof the plurality of symbols having the first symbol value and the lowsignal for each of the plurality of symbols having the second symbolvalue or the third symbol value for producing the first pulsed signaloutput; and generating the high signal for each of the plurality ofsymbols having the third symbol value and the low signal for each of theplurality of symbols having the second symbol value or the first symbolvalue for producing the second pulsed signal output comprises performingedge detection.